A Metamaterial-Based Compact Planar Monopole Antenna for Wi-Fi and UWB Applications

Ultrawideband (UWB) antennas are widely used as core devices in high-speed wireless communication. A novel compact UWB monopole antenna with an additional narrow band for Wi-Fi applications comprising a metamaterial (MTM) is proposed in this paper. The antenna has a compact size of 27 × 33 mm2 and consists of a V-shaped slot with two rectangular slots in the radiation patch. The inductance and capacitance develop due to the V-shaped slot in the radiation patch. The proposed antenna has −10 dB bandwidths of 3.2 GHz to 14 GHz for UWB and 2.38 GHz to 2.57 GHz for narrowband, corresponding to 144% and 7.66% fractional bandwidths, respectively. The measured gain and efficiency meet the desired values for UWB and Wi-Fi applications. To verify the performance of the antenna, the proposed antenna is fabricated and tested. The simulated and measured results agree well at UWB frequencies and Wi-Fi frequencies, and the antenna can be used as a smart device for portable IoT applications.


Introduction
Ultrawideband technology has promising prospects in short-range communication systems due to its attractive characteristics such as broad bandwidth, high data-rates, low power spectrum levels, and good radiation performance [1,2]. The UWB frequency range is defined as 3.1-10.6 GHz by the Federal Communications Commission (FCC) [3,4]. UWB antennas play a role in many applications, such as surveillance systems, medical applications, wireless sensor networks, and the Internet of Things (IoT) [5][6][7][8]. However, there are limitations on UWB antennas. These antennas are still large in size, and some frequency bands for short-range communications are not in the UWB spectrum. It would be advantageous to extend the scope of short-range communication systems and to meet commercial and functional requirements, which would require the addition of more frequency bands and a more compact size of UWB antennas for various services.
A number of approaches have been described in the literature for achieving UWB characteristics. In [9][10][11][12][13][14], the UWB technique was realized through split-ring resonators. Split-ring resonators can be used in many ways, such as for the etching of the patch or ground plane, embedded inside or onto the antenna substrate with different structures, and with a capacitive-loaded strip (CLS). Moreover, the analysis of these structures was always based on several factors, such as wider bandwidth, size, composition, and density. In [15][16][17][18][19], the UWB technique was implemented with a defective ground plane and different shapes of slots or stubs. Although some of these antennas were miniaturized in size, they did not integrate the UWB spectrum with other useful frequency bands for short-range communications.

Antenna Geometry
The presented antenna, as shown in Figure 1, exhibits dual-band operations with the help of antenna-loaded elements. The monopole antenna contains a 14.64 × 17 mm 2 radiation patch. The microstrip feedline length is set to 13.5 mm with a width of 2.3 mm to achieve 50 Ω characteristic impedance. The proposed antenna is designed on an FR-4 substrate with a dielectric constant of 4.4, and the tangent loss is δ = 0.02. In Figure 2, the V-slot is defined with an angle of α along with rectangular slots, and the realization of the optimum parameters is discussed in the parametric study.
The obtained values of the optimized patch are given in Table 1. There is a protracted area in the patch structure with the feedline, which helps to control the surface current. A stepped structure in the ground plane is used to extend the impedance bandwidth.

Composite Right Left Hand (CRLH) Metamaterial (MTM) Theory
To design a dual-band antenna for the Wi-Fi band and ultrawideband, we introduce inductive and capacitive elements by etching the radiation patch. These reactive elements introduce a phase element between the voltage and current, such as gap capacitance [25] and meander line inductance [29]. To obtain CRLH transmission, we etch the radiation patch, which produces capacitive effects.

Composite Right Left Hand (CRLH) Metamaterial (MTM) Theory
To design a dual-band antenna for the Wi-Fi band and ultrawideband, we introduce inductive and capacitive elements by etching the radiation patch. These reactive elements introduce a phase element between the voltage and current, such as gap capacitance [25] and meander line inductance [29]. To obtain CRLH transmission, we etch the radiation patch, which produces capacitive effects.

Composite Right Left Hand (CRLH) Metamaterial (MTM) Theory
To design a dual-band antenna for the Wi-Fi band and ultrawideband, we introduce inductive and capacitive elements by etching the radiation patch. These reactive elements introduce a phase element between the voltage and current, such as gap capacitance [25] and meander line inductance [29]. To obtain CRLH transmission, we etch the radiation patch, which produces capacitive effects. These effects cause backward waves that travel along the radiation patch to obtain the desired frequency resonant bands and improve the radiation performance along the direction of the radiation patch. The length and width of the slots play a vital role in controlling the magnitudes of the capacitance and inductance. To obtain the properties of the negative refractive index, left-hand effects are observed for a specific range of frequencies. In this novel structure, we observe the strong control of the resonance frequency of the Wi-Fi band and X-band due to CRLH. The CRLH behavior always works as an LC resonator or LC component [30,31]. The inductance and capacitance for series branch resonators and shunt branch resonators are given as follows: L = L R , C = C R , for series branch resonators, L = L L , C = C L , for shunt branch resonators, (1) where the subscripts L and R denote left-handed (LH) and right-handed (RH) possessions, respectively. The resonant frequency of the CRLH components for the LC circuit is determined by where ω L is the left-hand resonant frequency, and ω R is the right-hand resonant frequency.
To determine the dispersion relation of a CRLH transmission, we can use the following relationships: where β(ω) is the phase function of frequency, which can be purely real or purely imaginary depending on whether the radicand is positive or negative; ω se and ω sh denote series and shunt-branch resonators, respectively. The CRLH is increasingly dispersive as the frequency increases because the phase velocity (v p = ω/β) becomes increasingly dependent on the resonance frequency, which also shows that low frequencies dominate the LH while high frequencies dominate the RH. From Equation (3), we can identify whether the proposed model follows CRLH properties. In general, the series and shunt resonances of CRLH are different due to unbalanced cases because left-handed transmission supports electromagnetic waves with phase and group velocities that are antiparallel to each other. The apparent backward wave propagation is a key characteristic of CRLH transmission. Due to backward propagation, there is a certain shift in the transition frequency, called the CRLH gap. This gap is due to the left-handed effects. In the designed structure, the V-shaped resonator acts as a shunt capacitor and controls the voltage of the radiation patch to increase the power of the transmission line and reduce the antenna loss. If the bandgap for the left and right-hand region vanishes, then the CRLH region is in a balanced state. In a balanced state, the system impedance Z o , left-hand impedance Z L , and right-hand impedance Z R are equal. Using Equation (6), we can determine the left-hand and right-hand impedances. The characteristic impedance Z c is always frequency-dependent, as given in Equation (7), and can be related to the material intrinsic impedance η = µ/ε.
The characteristic impedance for balanced and unbalanced CRLH transmission is illustrated in Figure 3. In this figure, ω o is a transition frequency, as illustrated in Equation (8), where ω CL is the lower cut off frequency and ω CR is the higher cut off frequency. The transition frequency separates the left-handed region from the right-handed region. In determining the CRLH region, the permittivity ε and permeability µ values indicate a negative index for transition frequencies, as given in Equations (9) Sensors 2019, 19, 5426 5 of 13 and (10). If the index is positive, then the material is referred to as right-handed, and if the index is negative, then the material is referred to as left-handed.
The characteristic impedance for balanced and unbalanced CRLH transmission is illustrated in Figure 3. In this figure, ωo is a transition frequency, as illustrated in Equation (8), where ωCL is the lower cut off frequency and ωCR is the higher cut off frequency. The transition frequency separates the left-handed region from the right-handed region. In determining the CRLH region, the permittivity ε and permeability µ values indicate a negative index for transition frequencies, as given in Equations (9) and (10). If the index is positive, then the material is referred to as right-handed, and if the index is negative, then the material is referred to as left-handed.

Antenna Parametric Study
In this novel structure, we introduce a V-shaped resonator that gives us a narrow band for the desired Wi-Fi band, as shown in Figure 4. Furthermore, the V-shaped slot with extended slots shows us the CRLH characteristics, and by varying Gv1, we can adjust our narrow-band resonance frequency from 2.29 GHz to 2.75 GHz, as shown in Figure 5a. We observe that this parameter has a significant effect on the resonance frequency of the narrowband and has a small effect on the UWB

Antenna Parametric Study
In this novel structure, we introduce a V-shaped resonator that gives us a narrow band for the desired Wi-Fi band, as shown in Figure 4. Furthermore, the V-shaped slot with extended slots shows us the CRLH characteristics, and by varying Gv1, we can adjust our narrow-band resonance frequency from 2.29 GHz to 2.75 GHz, as shown in Figure 5a. We observe that this parameter has a significant effect on the resonance frequency of the narrowband and has a small effect on the UWB spectrum. Likewise, by varying Lv3, we observe the same frequency resonance change at the satellite band from 7.5 to 8.3 GHz, as shown in Figure 5b. These results are observed due to the unbalanced case of CRLH, where the CRLH gap is shifted from the right-hand region to the left-hand region. To verify this resonance deviation, we simulate both parameters together to observe the variation in the resonance frequency. By analyzing both parameters, we achieve a similar resonance response, as shown in Figure 5c. Except for slot parameters, other geometric parameters may also have effects on the antenna matching. Figure 6 show the effect of the substrate height, H, on the return loss when keeping all the other parameters fixed. It can be observed from Figure 6 that the substrate height has a significant effect on the impedance matching of high frequency near 11GHz. verify this resonance deviation, we simulate both parameters together to observe the variation in the resonance frequency. By analyzing both parameters, we achieve a similar resonance response, as shown in Figure 5c. Except for slot parameters, other geometric parameters may also have effects on the antenna matching. Figure 6 show the effect of the substrate height, H, on the return loss when keeping all the other parameters fixed. It can be observed from Figure 6 that the substrate height has a significant effect on the impedance matching of high frequency near 11GHz.   The surface current distributions at the frequencies of 2.48 GHz, 3.5 GHz, 9 GHz, and 11.4 GHz are described in Figure 7. The figure clearly shows that the current distributions are different for all four bands. In particular, when the antenna operates at 2.48 GHz, as shown in Figure 7a, the V-slot with an upper extended slot generates the 2.48 GHz frequency band. In Figure 7b, the current is distributed near the V-slot and between the rectangular slots on the radiation patch. In Figure 7c, the current is mainly distributed at the radiation patch corners and the lower part of the radiation patch at 9 GHz, and in Figure 7d, the current is concentrated on the lower part of the radiation patch and around the rectangular slots. These results show that the concentrations of current on the radiation patch are mainly located around the V-slot for lower frequencies, and for higher frequencies, the current concentrations are mainly located at the corners and bottom part of the radiation patch. spectrum. Likewise, by varying Lv3, we observe the same frequency resonance change at the satellite band from 7.5 to 8.3 GHz, as shown in Figure 5b. These results are observed due to the unbalanced case of CRLH, where the CRLH gap is shifted from the right-hand region to the left-hand region. To verify this resonance deviation, we simulate both parameters together to observe the variation in the resonance frequency. By analyzing both parameters, we achieve a similar resonance response, as shown in Figure 5c. Except for slot parameters, other geometric parameters may also have effects on the antenna matching. Figure 6 show the effect of the substrate height, H, on the return loss when keeping all the other parameters fixed. It can be observed from Figure 6 that the substrate height has a significant effect on the impedance matching of high frequency near 11GHz.     The surface current distributions at the frequencies of 2.48 GHz, 3.5 GHz, 9 GHz, and 11.4 GHz are described in Figure 7. The figure clearly shows that the current distributions are different for all    The surface current distributions at the frequencies of 2.48 GHz, 3.5 GHz, 9 GHz, and 11.4 GHz are described in Figure 7. The figure clearly shows that the current distributions are different for all  current is mainly distributed at the radiation patch corners and the lower part of the radiation patch at 9 GHz, and in Figure 7d, the current is concentrated on the lower part of the radiation patch and around the rectangular slots. These results show that the concentrations of current on the radiation patch are mainly located around the V-slot for lower frequencies, and for higher frequencies, the current concentrations are mainly located at the corners and bottom part of the radiation patch.

Experimental Results and Discussions
To verify the present antenna model, the proposed monopole MTM antenna is fabricated, as shown in Figure 8. Good agreements between the measurement and simulation S11 results are achieved, as shown in Figure 9. The small difference between the measured and simulated results may be caused by the fabrication tolerance and the SMA connector. The resultant impedance bandwidths of less than −10 dB are 2.38 to 2.57 GHz with a fractional bandwidth of 7.66% for narrowband and 3.2 to 14 GHz, which corresponds to a fractional bandwidth of 144%, for the UWB spectrum. We can observe the presence of the CRLH gap in the simulated and measured results.

Experimental Results and Discussions
To verify the present antenna model, the proposed monopole MTM antenna is fabricated, as shown in Figure 8. Good agreements between the measurement and simulation S 11 results are achieved, as shown in Figure 9. The small difference between the measured and simulated results may be caused by the fabrication tolerance and the SMA connector. The resultant impedance bandwidths of less than −10 dB are 2.38 to 2.57 GHz with a fractional bandwidth of 7.66% for narrowband and 3.2 to 14 GHz, which corresponds to a fractional bandwidth of 144%, for the UWB spectrum. We can observe the presence of the CRLH gap in the simulated and measured results. Figure 10 shows the measured realized gain and antenna efficiency. The realized gain is above 1 dB and the antenna efficiency is above 85% at 2.48 GHz, which is sufficient for short-range communication requirements. It is also observed that a large dip in the realized gain and efficiency occurs at 2.7 GHz. From 3.2 GHz to 14 GHz, the antenna gain is almost flat with a peak gain of 3.9 dB except for near the 8 GHz frequency band. Also, from 3.2 GHz to 14 GHz, the antenna efficiency is almost flat at 90% except for near the 8 GHz frequency band. and the antenna efficiency is above 85% at 2.48 GHz, which is sufficient for short-range communication requirements. It is also observed that a large dip in the realized gain and efficiency occurs at 2.7 GHz. From 3.2 GHz to 14 GHz, the antenna gain is almost flat with a peak gain of 3.9 dB except for near the 8 GHz frequency band. Also, from 3.2 GHz to 14 GHz, the antenna efficiency is almost flat at 90% except for near the 8 GHz frequency band.   communication requirements. It is also observed that a large dip in the realized gain and efficiency occurs at 2.7 GHz. From 3.2 GHz to 14 GHz, the antenna gain is almost flat with a peak gain of 3.9 dB except for near the 8 GHz frequency band. Also, from 3.2 GHz to 14 GHz, the antenna efficiency is almost flat at 90% except for near the 8 GHz frequency band.

Radiation Characteristics
The radiation characteristics of the proposed antenna are shown in Figure 11. The simulated and measured radiation characteristics for the E-plane (yz-plane) and the H-plane (xz-plane) of the antenna are analyzed at three different frequencies; i.e., 2.48, 7, and 11 GHz. There is a good agreement between the measured and simulated results for these frequencies, which indicates that the proposed antenna provides fairly good omnidirectional H-plane patterns and bidirectional E-plane patterns. Additionally, at higher frequencies, the high cross-polarization level is caused by the unequal phase distribution and the significant magnitude of higher-order modes at higher frequencies [23].

E-Plane
H-Plane

Radiation Characteristics
The radiation characteristics of the proposed antenna are shown in Figure 11. The simulated and measured radiation characteristics for the E-plane (yz-plane) and the H-plane (xz-plane) of the antenna are analyzed at three different frequencies; i.e., 2.48, 7, and 11 GHz. There is a good agreement between the measured and simulated results for these frequencies, which indicates that the proposed antenna provides fairly good omnidirectional H-plane patterns and bidirectional E-plane patterns. Additionally, at higher frequencies, the high cross-polarization level is caused by the unequal phase distribution and the significant magnitude of higher-order modes at higher frequencies [23].

Radiation Characteristics
The radiation characteristics of the proposed antenna are shown in Figure 11. The simulated and measured radiation characteristics for the E-plane (yz-plane) and the H-plane (xz-plane) of the antenna are analyzed at three different frequencies; i.e., 2.48, 7, and 11 GHz. There is a good agreement between the measured and simulated results for these frequencies, which indicates that the proposed antenna provides fairly good omnidirectional H-plane patterns and bidirectional E-plane patterns. Additionally, at higher frequencies, the high cross-polarization level is caused by the unequal phase distribution and the significant magnitude of higher-order modes at higher frequencies [23].

Comparison with Other Reported Designs
A comparison of the proposed monopole UWB antenna with other UWB antennas in terms of the design technique, size, peak gain, and operating band is shown in Table 2. It can be observed from the comparison table that our proposed antenna has some obvious advantages such as a simpler structure, a smaller size, and a wider UWB bandwidth with an extra narrowband.

Comparison with Other Reported Designs
A comparison of the proposed monopole UWB antenna with other UWB antennas in terms of the design technique, size, peak gain, and operating band is shown in Table 2. It can be observed from the comparison table that our proposed antenna has some obvious advantages such as a simpler structure, a smaller size, and a wider UWB bandwidth with an extra narrowband.

Conclusions
In this paper, a novel dual-band monopole antenna for Wi-Fi and UWB applications is proposed. A narrowband with the UWB band is introduced to combine different short-range communication applications in a single device. By etching and the inclusion of certain slots in the radiation patch, narrowband together with ultrawideband properties are achieved. The antenna is designed and fabricated, and the simulated response is correlated with the measured results. The measurement results indicate that the designed antenna can operate over 2.38 to 2.57 GHz for narrowband and 3.2 to 14 GHz for ultrawideband. The antenna has a nearly omnidirectional radiation pattern, with more than 90% antenna efficiency. The flat gain can meet short-range communication system requirements, which indicates that this antenna has significant potential for Wi-Fi and UWB applications. This antenna demonstrates that it is a good candidate for short-range applications, as this antenna can be easily integrated with portable devices incorporating multiple communication applications.
Author Contributions: A.K. proposed a novel antenna for UWB application. The antenna was optimized through simulation results and then fabricated. Measured results were correlated with simulation results in the writing of the manuscript. R.S. was the head of the research lab, and participated in data analysis. J.D. contributed to the measurement results and paper review.