Analysis and Implementation of a Phase-Shift Pulse-Width Modulation Converter with Auxiliary Winding Turns

: A phase-shift pulse-width modulation converter is studied and investigated for railway vehicle or solar cell power converter applications with wide voltage operation. For railway vehicle applications, input voltage range of dc converters is requested to have 30–40% voltage variation of the nominal input voltage. The nominal input voltages of dc converters on railway vehicles applications may be 37.5 V, 48 V, 72 V, 96 V and 110 V. Therefore, a new dc converter with wide input voltage operation from 25 to 150 V is presented to withstand di ﬀ erent nominal input voltage levels such as 37.5–110 V on railway power units. To realize wide input voltage operation, an auxiliary switch and auxiliary transformer windings are used on output side of conventional full-bridge converter to have di ﬀ erent voltage gains under di ﬀ erent input voltage values. Phase-shift pulse-width modulation is adopted in the developed dc converter to accomplish soft switching operation on power switches. To conﬁrm and validate the practicability of the presented converter, experiments based on a 300 W prototype were provided in this paper.


Introduction
High efficiency power converters have been developed in renewable energy power conversion. However, the output voltage of solar panel varies widely and the voltage value is related to solar intensity. Thus, dc converters with wide voltage operation are demanded to convert solar energy to electric power. The dc-dc converters are required on railway vehicles for communication systems, braking systems, electric door systems and solid state lighting systems. However, the nominal input voltages of these control systems are different. The normal input voltages for the different applications are 110 V, 72 V, 48 V and 37.5 V. The international standard EN50155 demands a minimum ±30% voltage variation of the nominal input voltage for dc-dc converters in railway applications. Therefore, high efficiency dc-dc converters with wide input voltage operation from 37.5 V to 110 V with ±30% voltage variation are necessary for railway systems. Two-stage converters [1][2][3] are normally employed to realize wide voltage operation. The buck, buck-boost or boost circuit topology can be used in the first stage and full-bridge, half-bridge, forward or flyback circuit topology can be used in the second stage. However, the drawbacks of two stage converters are low circuit efficiency and more circuit components. The dc converters with series/parallel connection to achieve wide input voltage operation have been presented in [4][5][6][7]. Full-bridge or half-bridge converters with phase-shift pulse-width modulation (PS-PWM) on the secondary side have been discussed in [8][9][10][11] to realize soft switching and wide voltage operation. Series resonant converters or PS-PWM converters with wide voltage operation have been proposed in [12][13][14][15][16][17][18][19]. However, the input or output voltage range proposed in circuit topologies [4][5][6][7][8][9][10][11][12][13][14][15][16][17][18][19] is less than 4:1, i.e. V in,max ≤ 4V in,min . For solar power conversion or

Circuit Structure
The proposed circuit with wide voltage operation is shown in Figure 1a. The basic structure of the studied circuit is a full-bridge circuit, Q1~Q4, Llk, and T, on input side and a center-tapped rectifier with two sets of secondary windings Ns1 and Ns2 and an auxiliary switch S on output side.   In relation to input voltage value, the studied circuit has two voltage operation ranges, low input voltage range and high input voltage range. When V in is at low voltage range V in = V in,min~3 V in,min (3:1 operation range), the auxiliary switch S is in the on-state (Figure 1b). D 2 and D 3 are reverse biased. The secondary winding turns N s1 + N s2 are connected to output side. The turn-ratio of transformer T is N p /(N s1 + N s2 ) and the converter has higher dc voltage gain for low input voltage operation. The effective duty cycle of full-bridge converter is depended on input voltage value to maintain output voltage constant. PS-PWM scheme is used to control phase-shift angle between the lagging-leg and leading-leg. Therefore, the soft switching characteristics of power switches on full-bridge converter can be achieved. If V in is at high voltage range V in = 2V in,min~6 V in,min (3:1 operation range), auxiliary switch S is in the off-state and D 1 and D 4 are off as shown in Figure 1c. The secondary turns N s1 are connected to output side and the transformer turn-ratio is N p /N s1 . Thus, the presented circuit has lower dc voltage gain in high input voltage operation. Based on the on/off states of auxiliary switch S and winding turns N s1 + N s2 or N s1 connected to output load, a soft switching full-bridge converter with 6:1 (V in,max /V in,min = 6) wide input voltage range is accomplished in the studied circuit.

Operation Principle
The duty cycle control is used to generate the proper switching signals and provide a stable output voltage against the variation of V in and I o . According to input voltage value, auxiliary switch S is controlled at on-state or off-state for low voltage range operation or high voltage range operation. In the studied converter, it is assumed that L m >> L lk , C Q1 = . . . = C Q4 = C oss and N s1 = N s2 = N s . The proposed circuit has two input voltage ranges, V in,min~3 V in,min and 2V in,min~6 V in,min . The pulse-width modulation waveforms and the equivalent circuits for two input voltage ranges are provided in Figures 2 and 3. For each input voltage range, the proposed circuit has ten modes in every switching cycle.

Low Voltage Range (S on)
When V in,min < V in < 3V in,min , auxiliary switch S is turned on. The effective duty cycle d eff of full-bridge circuit is related to input voltage. The transformer turn-ratio in low voltage range is n L = N p /(N s1 + N s2 ). The dc voltage gain is expressed as G L = V o /V in.L = 2d eff /n L where V in,L denotes V in in low input voltage range. The pulse-width modulation waveforms are given in Figure 2a  Mode 1 [t 0 , t 1 ]: In mode 1, Q 1 and Q 4 are on and v Lm ≈ V in due to L m >> L lk The power flow from input source to output load is through L lk , T, D 1 , L o and C o . The output inductor voltage v Lo ≈ V in /n L − V o > 0 and i Lo increases.
Mode 2 [t 1 , t 2 ]: Q 1 is off at time t 1 . Because of i p (t 1 ) > 0, C Q2 (C Q1 ) is discharged (charged). Due to C Q1 and C Q2 are low capacitances, i p and i D1 are almost constant. At time t 2 , v CQ2 = 0 and the time duration in mode 2 is calculated as ∆t mode 2 ≈ 2V in C oss n L /i Lo (t 1 ).
Mode 3 [t 2 , t 3 ]: Since i p (t 2 ) > 0 and v CQ2 (t 2 ) = 0, D Q2 conducts and Q 2 turns on after t 2 to have zero-voltage switching characteristic. In order to have soft switching operation, the dead time t d between Q 1 and Q 2 is needed and expressed as t d > ∆t mode 2 = 2V in C oss n L /i Lo (t 1 ). Since L lk << L m and R on,sw (turn-on resistance of switches Q 1 and Q 2 ) is about hundreds of milliohm, the transformer primary voltage v Lm and the secondary voltages v s,N1 and v s,N2 approximate zero voltage. D 1 and D 4 are forward biased. Then the inductor voltage v Lo = −V o and i Lo decreases. The primary current i P (t) ≈ i P (t 2 ) − V Q24,drop /r lk where r lk is the equivalent resistance on L lk and V Q24,drop is voltage drop on Q 2 and Q 4 .
Mode 4 [t 3 , t 4 ]: At time t 3 , S 4 is off. Because of i p (t 3 ) > 0, C S3 (C S4 ) is discharged (charged). Due to C S3 and C S4 are low capacitances, i p , i D1 and i D4 are constant. At time t 4 , v CQ3 = 0. The time duration in mode 4 is calculated as ∆t mode 4 ≈ 2V in C oss /i p (t 3 ).
Mode 5 [t 4 , t 5 ]: Because of i p (t 4 ) > 0 and v CQ3 (t 4 ) = 0, D Q3 is forward biased and S 3 turns on after t 4 to have zero-voltage switching operation. Due to both D 1 and D 4 are conducting, the primary magnetizing voltage v Lm is clamped at zero voltage. Thus, the leakage inductor voltage v Llk = −V in and i p is decreased. In this mode, i D1 (i D4 ) is decreased (increased) to 0 (i Lo ) at time t 5 . The time duration in mode 5 is expressed as: The duty loss in Mode 5 is calculated in (2): The waveforms in modes 6-10 are symmetry to waveforms in modes 1-5. Therefore, the analysis and discussion of modes 6-10 are neglected in this section. ip is decreased. In this mode, iD1 (iD4) is decreased (increased) to 0 (iLo) at time t5. The time duration in mode 5 is expressed as: The duty loss in Mode 5 is calculated in (2): The waveforms in modes 6-10 are symmetry to waveforms in modes 1-5. Therefore, the analysis and discussion of modes 6-10 are neglected in this section. (c)

High Voltage Range (S off)
When 2Vin,min < Vin ≤ 6Vin,min, S is off and D1 and D4 are reverse biased as shown in Figure 1c. Only the secondary turns Ns1 are connected to output load and the turn-ratio of transformer in high voltage range is nH = Np/Ns1. The effective duty cycle deff depends on Vin and Io. The dc voltage gain in high voltage range operation is given as GH = Vo/Vin,H = 2deff/nH where Vin,H denotes Vin in high input voltage range. Since turn-ratio nH = 2nL due to Ns1 = Ns2, it can obtain GH = GL/2 under the same effective duty cycle deff. That means the proposed converter has high voltage gain in low input voltage range and low voltage gain in high input voltage range. The circuit waveforms in high voltage range operation are provided in Figure 3a and the ten equivalent circuits for high voltage range operation are illustrated in Figure 3b-k. The circuit operation in Figure 3b-k with the secondary winding turns Ns1 in high voltage range are similar to the circuit operations in low voltage range presented in previous section. Therefore, the circuit analysis of the presented circuit operated at high voltage range are ignored in this section.

High Voltage Range (S off)
When 2V in,min < V in ≤ 6V in,min , S is off and D 1 and D 4 are reverse biased as shown in Figure 1c. Only the secondary turns N s1 are connected to output load and the turn-ratio of transformer in high voltage range is n H = N p /N s1 . The effective duty cycle d eff depends on V in and I o . The dc voltage gain in high voltage range operation is given as Since turn-ratio n H = 2n L due to N s1 = N s2 , it can obtain G H = G L /2 under the same effective duty cycle d eff . That means the proposed converter has high voltage gain in low input voltage range and low voltage gain in high input voltage range. The circuit waveforms in high voltage range operation are provided in Figure 3a and the ten equivalent circuits for high voltage range operation are illustrated in Figure 3b-k. The circuit operation in Figure 3b-k with the secondary winding turns N s1 in high voltage range are similar to the circuit operations in low voltage range presented in previous section. Therefore, the circuit analysis of the presented circuit operated at high voltage range are ignored in this section.

Circuit Characteristics and Design Example
Full-bridge PS-PWM converter with auxiliary secondary turns is adopted in the studied circuit to accomplish wide input voltage range operation. The effective duty cycle between the lagging-leg and leading leg is regulated under the different input voltage value and load current. For low (high) input voltage range, switch S is on (off) and the winding turns Ns1 + Ns2 (Ns1) are connected to output terminal. The output voltage can be derived in (3) from flux balance theory on Lo: where d is duty cycle of leg voltage vab and d = deff − dloss,5. The average value of output inductor current iLo equals Io. The voltage ratings of active switches Q1-Q4 equal Vin,max. The voltage ratings of D1-D4 and S are functions of Vin,max, nL and nH.
The average values of diode currents are ID1 = … = ID4 = Io/2 and IS,av = Io. The output inductance Lo approximates: The duty loss in Mode 5 is related to the leakage inductance Llk. If the maximum duty loss is defined, then the leakage inductance Llk is calc

Circuit Characteristics and Design Example
Full-bridge PS-PWM converter with auxiliary secondary turns is adopted in the studied circuit to accomplish wide input voltage range operation. The effective duty cycle between the lagging-leg and leading leg is regulated under the different input voltage value and load current. For low (high) input voltage range, switch S is on (off) and the winding turns N s1 + N s2 (N s1 ) are connected to output terminal. The output voltage can be derived in (3) from flux balance theory on L o : The average values of diode currents are I D1 = . . . = I D4 = I o /2 and I S,av = I o . The output inductance L o approximates: The duty loss in Mode 5 is related to the leakage inductance L lk . If the maximum duty loss is defined, then the leakage inductance L lk is calculated in (8): The proposed converter is designed with the following parameters: V in = 150-25 V (6:1 ratio), V o = 12 V, I o,rated = 25 A, P o = 300 W and f sw = 100 kHz. When V in = 25-75 V (low input voltage range), auxiliary switch S is on to obtain a high voltage gain due to the secondary turns N s1 + N s2 are connected to output side. If V in is increased and equal to 75 V, then switch S turns off and the converter is operated in high voltage range. The winding turns N s1 are connected to output side. If the input voltage is decreased from 150 V to 50 V, then switch S turns on and the circuit is operated in low voltage range. Thus, the proposed converter works in high voltage range when V in = 150-50 V. It is assumed the proposed converter has 85% efficiency at V in,min = 25 V and 100% power under low voltage range. The maximum duty cycle of leg voltage v ab at V in,min = 25 V is assumed 0.45 and the assumed duty cycle loss d loss,5 = 0.08: From (3) and (9), the leakage inductance approximates: From (3), the turn-ratio n L can be derived in (11): In the prototype circuit, transformer T is implemented using TDK (Tokyo Electric Chemical Industry Co., Ltd.) EER-42 ferrite core with winding turns N p = 12 and N s1 = N s1 = 4 and L m = 400 µH. The minimum effective duty cycle d eff,min at V in,L,max = 75 V is derived in (12): The maximum ripple current ∆i Lo at V in,L,max = 75 V (low input voltage range) is assumed 30% of load current. From (7), the inductance L o is expressed as: The L o = 8 µH is used in the prototype circuit. The rms (root-mean-square) currents of Q 1 -Q 4 are approximately calculated as: The maximum voltage on Q 1 -Q 4 is the maximum input voltage V in,max = 150 V. The average diode currents I D1 -I D4 are 12.5 A. The voltage ratings of D 1 -D 4 are given in (15) and (16): MOSFETs (metal-oxide-semiconductor field-effect transistor) IRFB4229 with 250 V/46 A ratings are adopted for active switch Q 1~Q4 and IRF3145 with 150 V/30 A ratings is adopted for auxiliary switch S. Fast recovery diodes APT30DQ60BG with 600 V/30 A ratings are selected for D 1 -D 4 and the output capacitance C o = 2000 µF. Figure 4a shows the control block of the signals Q 1~Q4 and S. Schmitt trigger comparator is adopted to generate the signal S in order to select the correct range of input voltage. The PWM signals are generated by phase-shift PWM integrated circuit UCC3895. The output voltage is controlled by the voltage regulator TL431 and PC817. Figure 4b provides the photograph of the prototype. Figure 5 demonstrates the simulated results of the primary-side and the secondary-side voltages and currents at full load under low input voltage range for both V in = 25 V and 75 V. One can observe that the effective duty cycle of v ab at V in = 25 V is larger than the duty cycle at V in = 75 V. Therefore, the current ripple ∆i Lo at V in = 25 V is less than current ripple ∆i Lo at V in = 75 V. Similarly, the simulated results of the presented circuit operated at high input voltage range are provided in Figure 6 for both V in = 150 V and 50 V. The lower input voltage (V in = 50 V) has the larger duty cycle to reduce the output ripple current ∆i Lo compared to the condition at V in = 150 V case. Experiments from a prototype with circuit components derived from previous section are provides to demonstrate the converter performance. The proposed circuit is operated at low input voltage range when V in is increased from 25 V to 75 V. Then, the circuit goes to the high input voltage mode. If the input voltage is decreased from 150 V to 50 V, then the circuit operation goes to low input voltage range. Based on the input voltage operation range. Figures 7-9 gives the test results when input voltage is in low voltage range. Under low input voltage range, auxiliary switch S is on. Thus, the rectifier diodes D 2 and D 3 are reverse biased and off. Figure 7a,b show the experimental waveforms of the gate voltages, leg voltage v ab and the primary current i p at V in = 25 V and 75 V and the rated output. It is clear that the converter has less phase-shift between Q 1 and Q 4 and larger duty cycle on leg voltage v ab at V in = 25 V input. Figure 7c,d provides the test waveforms of i D1 , i D4 and i Lo at V in = 25 V and 75 V and the rated power. Due the converter at V in = 25 V input has large duty cycle, i Lo has less current ripple at V in = 25 V input compared to V in = 75 V input. Figure 8a,b provides the experimental waveforms of switch Q 1 in leading-leg at 20% and 100% loads under V in = 25 V input. It is clear the soft switching operation of Q 1 is achieved from 20% load under 25 V input case. In the same manner, test waveforms of Q 1 at 20% and 100% loads under 75 V input are given in Figure 8c,d. It can observe that Q 1 is also turned on at zero-voltage switching from 20% load at V in =75 V input case. Switches Q 1 and Q 2 have the same turn-on/off characteristics in the leading-leg. Therefore, the soft switching operation of switches Q 1 and Q 2 at leading-leg can be accomplished from 20% load in low input voltage range operation. Figure 9a,b show the test waveforms of the switch Q 3 at lagging-leg for 20% and 100% loads under 25 V input. It is clear that Q 3 is also turned on at zero voltage switching from 20% load. Figure 9c,d provide the measured waveforms of Q 3 at 75 V input. It can observe that Q 3 is hard switching at 75 V input case for both 20% and 100% loads. The soft switching condition of power switches at lagging-leg (leading-leg) is related to the leakage inductor (output inductor). Normally, the leakage inductance is much less than the output inductance. Therefore, power switches at leading-leg have wide zero-voltage switching range compared to the lagging-leg switches. The test results shown in Figures 8 and 9 agree with these statements. Figures 10-12 gives the experimental waveforms of the proposed converter for high input voltage range operation at 150 V and 50 V input. The test waveforms of v Q1,g , v Q4,g , v ab and i p at V in = 150 V and 50 V under rated power are provided in Figure 10a,b. Compared to 150 V input case, the proposed converter has less phase-shift between Q 1 and Q 4 and larger duty cycle on leg voltage v ab at V in = 50 V input. The test results of i D1 , i D4 and i Lo at 100% load under 150 V and 50 V input are shown in Figure 10c,d. Figure 11 provides the experimental waveforms of leading-leg switch Q 1 at various load power and input voltage values in high voltage range operation. It is clear that the soft switching operation of Q 1 is accomplished from 20% load for both 150 V and 50 V input. Similarly, the test waveforms of lagging-leg switch Q 3 at various load power and input voltage values are demonstrated in Figure 12. It can be observed that Q 3 has soft switching operation only at 50 V input under 100% load and hard switching operation for the other conditions. Figure 13 provides the test results of the circuit efficiencies. For both low and high voltage range operation, the lower input voltage has larger effective duty cycle and less conduction losses on power switches compared to the higher input voltage. Thus, the circuit efficiency at 25 V input is better than 75 V input in low input voltage range. Similarly, the circuit efficiency at 50 V is better than 150 V input in high voltage range operation.

Simulated and Experimental Results
voltage range. Similarly, the circuit efficiency at 50 V is better than 150 V input in high voltage range operation.

Conclusions
A full-bridge PS-PWM converter with auxiliary secondary turns is investigated to demonstrate the capability of 6:1 wide input voltage range operation for railway vehicle low power units and solar power converters. To achieve wide voltage range operation, two sets of secondary windings are used in the studied circuit. According to the different secondary winding turns connected to output terminal, there are two input voltage range operations in the converter. In each voltage range, the circuit can realize the 3:1 input voltage range operation. For low voltage range, the high transformer turns-ratio is adopted to achieve higher dc voltage gain. On the other hand, a low turns-ratio of transformer is used to obtain lower dc voltage gain of the studied circuit. Therefore, a 6:1 wide voltage operation is implemented in the presented converter. PS-PWM scheme is used to accomplish soft switching operation for leading-leg switches. The feasibility of the studied converter is verified and confirmed by a design example and some experiments in this paper.

Conclusions
A full-bridge PS-PWM converter with auxiliary secondary turns is investigated to demonstrate the capability of 6:1 wide input voltage range operation for railway vehicle low power units and solar power converters. To achieve wide voltage range operation, two sets of secondary windings are used in the studied circuit. According to the different secondary winding turns connected to output terminal, there are two input voltage range operations in the converter. In each voltage range, the circuit can realize the 3:1 input voltage range operation. For low voltage range, the high transformer turns-ratio is adopted to achieve higher dc voltage gain. On the other hand, a low turns-ratio of transformer is used to obtain lower dc voltage gain of the studied circuit. Therefore, a 6:1 wide voltage operation is implemented in the presented converter. PS-PWM scheme is used to accomplish soft switching operation for leading-leg switches. The feasibility of the studied converter is verified and confirmed by a design example and some experiments in this paper.

Conclusions
A full-bridge PS-PWM converter with auxiliary secondary turns is investigated to demonstrate the capability of 6:1 wide input voltage range operation for railway vehicle low power units and solar power converters. To achieve wide voltage range operation, two sets of secondary windings are used in the studied circuit. According to the different secondary winding turns connected to output terminal, there are two input voltage range operations in the converter. In each voltage range, the circuit can realize the 3:1 input voltage range operation. For low voltage range, the high transformer turns-ratio is adopted to achieve higher dc voltage gain. On the other hand, a low turns-ratio of transformer is used to obtain lower dc voltage gain of the studied circuit. Therefore, a 6:1 wide voltage operation is implemented in the presented converter. PS-PWM scheme is used to accomplish soft switching operation for leading-leg switches. The feasibility of the studied converter is verified and confirmed by a design example and some experiments in this paper.