Wireless THz link with optoelectronic transmitter and receiver

Photonics might play a key role in future wireless communication systems that operate at THz carrier frequencies. A prime example is the generation of THz data streams by mixing optical signals in high-speed photodetectors. Over the previous years, this concept has enabled a series of wireless transmission experiments at record-high data rates. Reception of THz signals in these experiments, however, still relied on electronic circuits. In this paper, we show that wireless THz receivers can also greatly benefit from optoelectronic signal processing techniques, in particular when carrier frequencies beyond 0.1 THz and wideband tunability over more than an octave is required. Our approach relies on a high-speed photoconductor and a photonic local oscillator for optoelectronic down-conversion of THz data signals to an intermediate frequency band that is easily accessible by conventional microelectronics. By tuning the frequency of the photonic local oscillator, we can cover a wide range of carrier frequencies between 0.03 THz and 0.34 THz. We demonstrate line rates of up to 10 Gbit/s on a single channel and up to 30 Gbit/s on multiple channels over a distance of 58 m. To the best of our knowledge, our experiments represent the first demonstration of a THz transmission link that exploits optoelectronic signal processing techniques both at the transmitter and the receiver.


Introduction and background
Data traffic in wireless communication networks is currently doubling every 22 months 1 and will account for more than 60 % of the overall internet traffic by 2021. Sustaining this growth requires advanced network architectures that combine massive deployment of small radio cells [2][3][4] with powerful backhaul infrastructures, built from high-capacity wireless point-to-point links. Such links may be efficiently realized by exploiting THz carriers in low-loss atmospheric transmission windows 5 , thereby offering data rates of tens or even hundreds of Gbit/s. To generate the underlying communication signals at the THz transmitter, optoelectronic signal processing [6][7][8][9][10][11] has emerged as a particularly promising approach, leading to demonstrations of wireless transmission at line rates of 100 Gbit/s and beyond [12][13][14][15][16][17][18] . At the THz receiver, however, the advantages of optoelectronic signal processing have not yet been exploited.
In this paper, we show that wireless THz receivers can benefit from optoelectronic signal processing techniques as well, in particular when carrier frequencies beyond 0.1 THz and wideband tunability are required 19,20 . We exploit a high-speed photoconductor and a photonic local oscillator for terahertzto-electrical down-conversion over a broad range of frequencies between 0.03 THz and 0.34 THz. In our experiments, we demonstrate a coherent wireless link that operates at a carrier frequency of 0.31 THz and allows line rates of up to 10 Gbit/s on a single channel and up to 30 Gbit/s on multiple channels over a distance of 58 m. To the best of our knowledge, this represents the first demonstration of a THz transmission link that complements optoelectronic signal generation at the transmitter by optoelectronic down-conversion at the receiver.
The vision of a future wireless network architecture is shown in Fig. 1(a). The increasing number of terminal devices and the advent of new data-hungry applications require a dense mesh of small radio cells to provide ubiquitous broadband wireless access [2][3][4] . Backhauling of these cells relies on high-speed wireless point-to-point links, which are seamlessly integrated into fibre optical networks 21,22 . The high data rates required for wireless backhauling infrastructures are achieved by using carrier frequencies in the range of 0.1 THz to 1 THz (T-waves). Figure 1(b) shows the atmospheric Twave attenuation as a function of frequency 23 , revealing several transmission windows with low attenuation that can be used for wireless communications. For highest flexibility and performance, T-wave transmitters (Tx) and receivers (Rx) should be able to switch between various windows depending on channel occupancy and weather conditions. At the Tx, this can be achieved by optoelectronic T-wave signal generation, Fig. 1(c), which relies on mixing of an optical data signal at a carrier frequency S,a f with continuous-wave (c.w.) tone at frequency S,b f in a high-speed photodiode (optical-to-Twave, O/T conversion). This leads to a T-wave data signal centred at the carrier frequency SS , a S , b f ff =−, which can be widely tuned by changing the frequency S,b f of the unmodulated optical tone. Note that similar optoelectronic Tx concepts have been used in earlier demonstrations 12-18 , but were complemented by an electronic Rx which cannot match the wideband tunability of the Tx. To overcome this limitation, we have implemented an optoelectronic Rx that requires neither an electronically generated local oscillator (LO) nor an electronic mixer for coherent reception, but relies on a photoconductor that is driven by a photonic LO instead, Fig. 1d. The photonic LO is generated by superimposing two optical c.w. tones with frequencies LO,a f and LO,b f and is coupled to the photoconductor 24-26 for down-conversion of the T-wave data signal to an intermediate frequency band that is easily accessible by conventional microelectronics (T-wave-to-electrical, T/E conversion).
Note that optically driven photoconductors have previously been used for down-conversion of THz waves in spectroscopy systems 24-28 . These demonstrations, however, are usually based on THz Tx and Rx that are driven by a common pair of lasers ( S,a LO,a ff = , S,b LO,b ff = ) for homodyne reception, and they rely on narrowband detection schemes with typical averaging times of the order of 1 ms and bandwidths of a few kHz for highly sensitive acquisition of small power levels. In our work, we advance these concepts to enable wireless data transmission at GHz bandwidth over technically relevant distances, using the complex amplitude of the THz wave for encoding of information. To this end, we make use of advanced photoconductors with engineered carrier lifetime 24 , we combine them with high-speed transimpedance amplifiers 29 into a perfectly shielded read-out circuit, and we exploit heterodyne detection in combination with advanced digital signal Fig. 1. T-wave wireless infrastructure using optoelectronic techniques. (a) Vision of a future wireless network architecture. A dense mesh of small radio cells provides broadband wireless access to a vast number of users and devices. The high data rates required for the underlying wireless backhauling infrastructures are provided by high-speed wireless point-to-point links that are operated at THz frequencies and that can be efficiently interfaced with fibre-optic networks. (b) T-wave atmospheric attenuation for standard conditions 23 (temperature of 15 °C, water-vapour content of 7.5 g/m 3 ). Various windows with low attenuation can be used for T-wave communications. Our Rx allows operation over a wide range of frequencies between 0.03 THz and 0.34 THz, in which the atmospheric attenuation is small enough to permit transmission over technically relevant distances. (c) Optoelectronic T-wave signal generation. The data signal is modulated on an optical continuous-wave (c.w.) tone with frequency S,a f by an electro-optic modulator (EO mod). The modulated signal is superimposed with an unmodulated c.w. tone S,b f . The optical signal is converted to a T-wave signal in a high-speed photodiode (optical-to-T-wave conversion, O/T) which is radiated into free space by an antenna. The carrier frequency of the T-wave is given by the frequency difference processing to overcome phase noise and drift associated with the free-running photonic LO at the Rx. In the following, we give details about the optoelectronic Rx module and the demonstration of the T-wave link.

Implementation of optoelectronic receiver
The concept and the implementation of the optoelectronic Rx is illustrated in Fig. 2 The free carriers generated by the absorbed optical power change the photoconductance according to where G denotes a proportionality constant that describes the sensitivity of the photoconductor. The resulting current () This intermediate signal contains the amplitude and phase information of the T-wave data signal. A more detailed derivation of Eqs. (1)-(4) can be found in Supplementary Section 1. Figure 2(b) illustrates the technical implementation of the Rx module used for our experiments. The photoconductor 24 (PC) is connected to a bow-tie antenna which is electrically coupled to a TIA by a metal wire bond. The photoconductor is operated without any DC bias, and a decoupling capacitor 1nF C = is used to isolate the device from the bias that is effective at the input of the TIA, see Supplementary Section 2. The output of the TIA is electrically connected to a printed circuit board. The photoconductor is illuminated from the top with the optical power () LO Pt , which is coupled from the horizontally positioned fibre by a photonic wire bond 30 . The entire assembly is glued on a silicon lens which focuses the incoming T-wave to the antenna, see Fig. 2(c), and the assembly is placed in metal housing for effective electromagnetic shielding. A microscope image of the fabricated device and a more detailed description of the Rx module in terms of conversion efficiency, bandwidth and noise can be found in Supplementary Section 2.

Demonstration of wireless THz links
In the following we demonstrate the viability of our receiver concept in a series of experiments, covering both single-channel and multi-channel transmission of THz data signals. For the single-channel experiments, our focus was on demonstrating the wideband tunability of the carrier frequency. The multi-channel experiment shows the scalability of the approach towards high-throughput parallel transmission.

Single-channel transmission and wideband tunability
The wireless transmission system for single-channel transmission is illustrated in Fig. 3(a). As data source we use an arbitrary-waveform generator (AWG) which provides a quadrature phase shift keying (QPSK) data to an optical IQ-modulator. The modulator is fed by a tunable laser with an optical carrier frequency S,a f . The modulated data signal is superimposed by an unmodulated optical carrier with frequency S,b .
f The optical power spectrum of the data signal and the unmodulated laser tone for a 3GBd QPSK signal and a frequency spacing of SS , a S , b 0.310 THz ff f =−= is shown in Inset 1 of Fig. 3(a). For O/T-conversion, we use a commercially available uni-travelling-carrier photodiode 31 (UTC-PD). The T-wave signal is then radiated into free space by a horn antenna and transmitted over a distance of 58 m, limited only by the size of our building. At the Rx, we use a horn antenna to couple the T-wave signal into a WR 3.4 hollow waveguide that is connected to a two-stage T-wave amplifier 32 which compensates the transmission loss. At the output of the amplifier another horn antenna is used to feed the signal to the silicon lens of the T-wave Rx. T/E-conversion is then performed in the photoconductor which is illuminated by a photonic LO, see power spectrum in Inset 2 of Fig. 3(a). A comprehensive description of the transmission setup and a characterization of the T-wave amplifiers and the UTC-PD is given in Supplementary Sections 3 and 4.
After down-conversion, the electrical signals are recorded by a real-time oscilloscope and stored for offline digital signal processing (DSP). The T-wave Rx relies on a heterodyne de- where the photonic LO is placed at the edge of the T-wave data spectrum, which leads to electrical signals centred around the intermediate frequency The in-phase and the quadrature components of the QPSK baseband signals are then extracted from the intermediate signals by DSP, comprising standard procedures such as digital frequency down-conversion, timing recovery, constant-modulus equalization, frequency offset compensation, and carrier phase estimation. Figure 3(b) shows the bit error ratio (BER) measured for various line rates b R at a carrier frequency of 0.31 THz. For line rates below 3 Gbit/s, no errors are measured in our recording length of 10 5 symbols, demonstrating the excellent performance of the optoelectronic Rx. The constellation diagrams for line rates of 1.5 Gbit/s, 5 Gbit/s and 10 Gbit/s are shown in the insets. For larger line rates, the received signal quality decreases mainly due to limitations of the TIA in the intermediate-frequency circuit. The TIA has a specified bandwidth of only 1.4 GHz and larger line rates would require a more broadband device. With the current TIA, we could transmit 10 Gbit/s with a BER below the threshold of forward-error correction (FEC) with 7 % overhead. Note that the transmission distance of 58 m was dictated by space limitations. With the current components, it would be possible to bridge roughly twice the distance by doubling the optical power at the Tx.
To the best of our knowledge, our experiments represent the first demonstration of a THz transmission link that complements optoelectronic generation of T-wave signals at the Tx by opto-electronic down-conversion at the Rx. Similar schemes have also been demonstrated 33,34 at lower carrier frequencies, relying on a UTC-PD for optoelectronic T/E conversion. Using this Rx, a line rate of 5 Gbit/s at a carrier frequency of 35.1 GHz and a line rate of 1 Gbit/s at a carrier frequency of 60 GHz have been demonstrated with transmission distances of 1.3 m and 0.55 m, respectively. Our work relies To further demonstrate the flexibility of the optoelectronic Rx, we transmit 2 Gbit/s data streams at various carrier frequencies covering the entire range between 0.03 THz and 0.34 THz. Note that the setup shown in Fig. 3(a) only allows to cover the frequency range between 0.24 THz and 0.34 THz due to bandwidth limitations of both the UTC-PD and the Twave amplifiers. For transmission at frequencies between 0.03 THz and 0.18 THz, we omitted the amplifiers and replaced the UTC-PD by a pin-PD. The measured BER and some exemplary constellation diagrams of the transmission experiments are shown in Fig. 4. For carrier frequencies between 0.24 THz and 0.34 THz, no errors were measured in our recordings such that we can only specify an upper limit of 10 -4 for the BER. For carrier frequencies between 0.03 THz and 0.18 THz, comparable performance was obtained. For simplicity, the transmission experiments in the lower frequency range were performed over a transmission distance of 7 cm only, which could be bridged without any amplifiers. The range could be easily extended to tens or hundreds of meters by using Rx antennae that are optimized for lower frequencies in combination with amplifiers. Note also that the data points between 0.03 THz and 0.18 THz were taken on a slightly irregular frequency grid, thereby avoiding some carrier frequencies for which the free-space link of our setup features low transmission. This does not represent a fundamental problem, but was caused by fading due to uncontrolled reflections in the beam path. The associated power variations could be overcome by using amplifiers with adaptive gain or by optimizing the beam path for each frequency point individually. This fading in combination with frequency-dependent Tx power and a decreasing gain of the on-chip bow-tie antenna for low frequencies is also the reason for the degraded performance of the transmission experiments at carrier frequencies of 0.03 THz, 0.04 THz, 0.10 THz, and 0.18 THz. Still, these experiments demonstrate that the same receiver concept as in Fig. 2 can be used for carrier frequencies in a range of S 0.03THz 0.340 THz f ≤≤ , i.e. over more than a decade.

Multi-channel transmission
We also investigate the receiver in a multicarrier transmission experiment at carrier frequencies between 0.287 THz and 0.325 THz. We simultaneously transmit up to 20 T-wave channels (Ch 1 … 20) spaced by 2 GHz, where each channel is operated with a QPSK signal at a symbol rate (line rate) of 0.75 GBd (1.5 Gbit/s). To keep the experimental setup simple, we use a single broadband AWG and a single IQ-modulator to generate an optical signal that simultaneously contains all channels, which is then converted to the THz range by a UTC-PD. This approach allows us to re-use the experimental setup shown on Fig. 3(a). Alternatively, multiple optical carriers and less broadband devices in combination with optical multiplexing could have been used to generate the optical channels 14 . Figure 5 It is also worth mentioning that the 0.75 GBd T-wave channels were transmitted on a 2 GHz grid to avoid interference of data signals from neighbouring channels after down-conversion to the intermediate frequency band. This leads to un-used spectral regions of approximately 1.2 GHz between the T-wave channels, which could be avoided by optoelectronic down-conversion schemes that allow simultaneous extraction of the inphase and the quadrature component of the T-wave signal. Figure 5(c) shows the BER for transmission experiments with 6, 12, and 20 channels. For transmission of 12 channels (aggregate line rate 18 Gbit/s), the BER stays below the 7% FEC limit, whereas for 20 channels (30 Gbit/s), 20 % FEC overhead is required. In Fig. 5(d), the constellation diagrams of the 20 channel experiment are displayed. Note that in our current experiments, the data per channel was only limited by the bandwidth of the TIA. Using more broadband devices 35 , symbol rates of more than 25 GBd and data rates of more than , a waveguide-coupled uni-travelling-carrier photodiode (UTC-PD) is used in combination with a cascade of two T-wave amplifiers, as described in Fig. 3. For both cases, the same Rx module as in Fig. 2 was used, demonstrating its large tunability.

Summary
In summary, we showed a first demonstration of a coherent wireless THz communication system using optoelectronic signal processing both at the transmitter and at the receiver. Our experiments show that the same receiver concept can be used over a broad frequency range S 0.03THz 0.340 THz f ≤≤ , spanning more than a decade. We transmit a line rate of 10 Gbit/s using a single T-wave channel at a carrier frequency of 0.31 THz with a BER below the 7% FEC limit. In this experiment, the line rate was limited by the bandwidth of the transimpedance amplifier, but not by the transmitter and receiver scheme. We further demonstrate multi-channel transmission using up to 20 carriers with frequencies in the range between 0.287 THz and 0.325 THz. This leads to an aggregate line rate of 30 Gbit/s with a BER below the threshold for a FEC with 20 % overhead. The single and the multi-channel T-wave link bridges a distance of 58 m. In the future, optoelectronic T-wave receivers may exploit integrated frequency converters that may be efficiently realized on the silicon plasmonic platform 9 . Our findings demonstrate that coherent T-wave receivers with an optoelectronic, widely tunable local oscillator may build the base for a novel class of THz communication systems.

Detailed derivation of formulae
In the main paper, we show coherent wireless THz communication using an optoelectronic receiver 1-3 and a tunable photonic local oscillator (LO). The concept of optoelectronic down-conversion in a photoconductive T-wave receiver (Rx) is illustrated in Fig. 2(a) of the main paper. In the following, we give a detailed derivation of the associated formulae.
The T-wave data signal from the transmitter (Tx) at an angular carrier frequency SS π 2 f The photocarriers generated by the absorbed optical power change the photoconductance according to where G denotes a proportionality constant that describes the sensitivity of the photoconductor. The resulting current () The intermediate signal contains the amplitude and phase information of the T-wave data signal and can be processed by low-frequency electronics.

T-wave receiver
This section gives details of the implementation and the characterization of the optoelectronic Rx used in our experiments. Figure S1 shows images of our Rx module. The photoconductor 2 (PC) is in direct contact with a bow-tie antenna, see Fig. S1(a). The antenna is electrically bonded to a transimpedance amplifier (TIA, Maxim Integrated 4 PHY1097) for processing the down-converted intermediate-frequency current. The TIA is designed for amplification of receiver signals in a passive optical network, where the photodiode is reversed biased by the TIA. In our application, the photoconductor does not require a bias voltage and hence we use a capacitor C to decouple the photoconductor from the bias at the TIA input terminals. The output of the TIA is electrically bonded to a printed circuit board (PCB) having a gold-plated alumina ceramic substrate. The photoconductor is illuminated from the top with the time-dependent optical power ()

LO
Pt , which is coupled to the active region of the device from the horizontally positioned fiber by a photonic wire bond 5,6 , see inset of Fig. S1(a). The assembly is placed on a silicon lens which focuses the incoming T-wave onto the antenna, see Fig. 2(c) of the main manuscript. All components are placed in a metal housing for electromagnetic shielding of the Rx circuits and for simplified handing of the Rx. The fully packaged Rx is shown in Fig. S2(b). The photonic LO is fed to the Rx with a fiber, and the down-converted RF data signal ("data out") is processed further by standard laboratory equipment.
In the following, we give a detailed characterization of the Rx in terms of conversion efficiency, bandwidth, and noise.

Conversion efficiency
First we quantify the frequency-dependent response of the photoconductor connected to a bow-tie antenna. We define the conversion efficiency η as the ratio of the output power (S8) Figure S2 shows the conversion efficiency of the photoconductive Rx in dependence of the frequency S f . The grey hatched area indicates the frequency range used in our experiments. In general, the T-wave bandwidth of a photoconductor is limited by the lifetime τ of the free carriers that are generated by the incident optical signal. This lifetime can be reduced by low-temperature growth of the associated III-V materials 2 . The frequency response of our device shows a rolloff of the conversion efficiency for frequencies beyond   [ ] S THz f waveguide by a horn antenna. The waveguide is connected to the input of two cascaded T-wave amplifiers 10 , which compensate the free-space transmission loss and amplify the Twave. In our current design, we use another horn antenna at the output of the second T-wave amplifier in combination with a silicon lens to couple the T-wave to the photoconductor. In the future, the performance of the scheme may be further improved by replacing this assembly with a waveguidecoupled photoconductor. For generating the photonic LO, two c.w. laser tones with optical frequencies LO,a f and LO,b f are superimposed using a polarization-maintaining 50/50 coupler, thus generating an optical power beat. The beat signal is amplified by an EDFA followed by a 3 nm filter to reduce ASE noise. A polarization controller is used to maximize the IF signal at the output of the polarization-sensitive photoconductor. With an attenuator we adjust the optical power LO P at the Rx. Finally, the down-converted IF signal is coupled to the TIA, the output voltage of which is sampled and stored in a real-time oscilloscope for further offline signal processing. Figure S5(b) shows a photograph of the wireless transmission link. The image on the left shows the Tx including the UTC-PD and the T-wave PFTE lens. The Rx is 58 m away from the Tx. On the right image, the Rx including the T-wave PTFE lens, the T-wave amplifiers and the Rx module is shown in more detail. To facilitate identification of the components shown in the setup sketch of Fig. S5(a), we mark them with the letters A -H.
For finding optimum operation parameters, we characterize the performance of the wireless link shown in Fig. S5  at the Rx. For some measurement points, the signal quality is so high that we could not measure any errors in a recording length of 10 5 symbols. We therefore estimate the BER from the error vector magnitude 11 (blue dots). For an optical power of S,1 8mW P > , the signal quality decreases because the T-wave amplifiers saturate, see constellation diagrams in the right-hand side column of Fig. S6(a). For high optical powers S, 1 P , saturation of the Twave amplifiers leads to an asymmetric distribution of the noise around the various constellation points, whereas a symmetric distribution is observed for low optical powers.  , close to its optimum point shown in Fig. S6(a). The signal quality improves with increasing optical LO power LO,1 P and is finally limited by the maximum optical power that the photoconductor can withstand. 4. T-wave amplifiers and UTC-PD To compensate free space T-wave transmission loss, we use a cascade of a low-noise amplifier (LNA) and a medium-power amplifier (MPA), designed for operation in the submillimeter H-band (0.220 THz -0.325 THz), see Fig. S7(a). The moduli of the S-parameters for this cascade are shown in Fig. S7(b). In a frequency range from 0.260 THz to 0.335 THz, the total gain is more than 40 dB. To measure the frequency response of T-wave components and of the complete transmission system, we use the setups shown in Fig. S8. Two unmodulated c.w. laser tones having equal powers and different frequencies S,a f and S,b f are superimposed in a 50/50 combiner and coupled to the UTC-PD.
The T-wave output power THz P is measured in a calorimeter (VDI, Erickson PM4). By tuning the difference frequency SS , a S , b f ff =− of the two lasers we measured the frequency dependent output power of the UTC-PD without any amplifier, with the MPA, or with the cascade of LNA and MPA, Fig. S8(a,b,c). Furthermore, we measured the power after Twave transmission over 58 m with the cascaded LNA and MPA at the Rx, Fig. S8(d). The results of all these measurements are shown in Fig. 8(e,f). Note that optical input power of the UTC-PD had to be strongly reduced for the case of the cascaded T-wave amplifiers without free-space link to prevent amplifier saturation. The received THz power after the amplifier cascade is more than 0 dBm in a frequency range from 0.29 THz to 0.33 THz for a transmission distance of 58 m. P Red dots denote values that were directly measured, whereas blue dots refer to BER values estimated from the respective error vector magnitude (EVM). Since the length of our signal recordings was limited to 10 5 symbols, the lowest statistically reliable measured BER amounts to 10 -4 . For measured BER values above this threshold, directly measured and estimated BER show good agreement, giving us confidence that the EVM-based estimations for BER < 10 -4 are valid. (b) BER as a function of the LO power amplitude LO,1 P .