U-Shaped PN Junctions for Efficient Silicon Mach-Zehnder and Microring Modulators in the O-Band

We demonstrate U-shaped silicon PN junctions for energy efficient Mach-Zehnder modulators and ring modulators in the O-band. This type of junction has an improved modulation efficiency compared to existing PN junction geometries, has low losses, and supports high-speed operation. The U-shaped junctions were fabricated in an 8” silicon photonics platform, and they were incorporated in travelling-wave Mach-Zehnder modulators and microring modulators. For Mach-Zehnder modulators, the DC VπL at -0.5 V bias was 4.6 V·mm, and a VπL as low as ~2.6 V·mm was measured. A 2-mm long device had a 3dB bandwidth of 13 GHz and supported 24 Gb/s modulation. The ring modulator tuning efficiency was 40 pm·V between 0 V and -0.5 V bias. It had a 3dB bandwidth of 13.5 GHz and supported 13 Gb/s modulation. 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Introduction
Silicon (Si) optical modulators, in the form of Mach-Zehnder modulators (MZMs) and ring modulators, are an attractive solution for high-bandwidth electrical-to-optical conversion, since they can be fabricated at the wafer-scale in foundry processes [1][2][3][4][5][6][7][8].These devices typically use carrier accumulation and depletion via the plasma dispersion effect, which is weak and is lower in the O-band than the C-band [1].An increased modulation efficiency reduces the device length and drive voltage, which reduces the power consumption if the insertion loss (IL) is not compromised [9].
Si modulation phase-shifters can be formed using PN junction or silicon-insulator-silicon capacitor (SISCAP) structures.Lateral PN junctions are the most common, and they have a relatively high V π L around 2.5 V•cm, but low optical loss of around 10 dB/cm [2,3].Travelling-wave electrode designs have been demonstrated for lateral junction to support high-speed operation [10][11][12][13][14][15], up to 41 GHz [16].Vertical junctions [17][18][19][20] have V π L of 0.75 V•cm in the C-band, but they tend to cause high waveguide losses of about 31 dB/cm and their high capacitance limits the bit rates in MZMs to 16 Gb/s so far [20].Ring modulators with vertical junctions have been demonstrated to support 25Gb/s operation [19].Interdigitated junctions [21][22][23] have V π L of 1.12 V•cm in the C-band, but they also have high losses of 25 dB/cm and have worked at up to 10 Gb/s [22].A higher bit rate (40 Gb/s) has also been demonstrated with interdigitated junctions for which the V π L is 1.5-2.0V•cm and the loss is around 10 dB/cm [23].25Gb/s operation has been shown for the ring modulators with interdigitated junction [24].In the O-band, to date, the highest efficiency Si MZMs use carrier accumulation in the SISCAP geometry and have a V π L of 0.2 V•cm, but the propagation loss is very high at 65 dB/cm [8].
In this article, we present the design and measurements of U-shaped PN junctions for efficient Si MZMs and ring modulators operating in the O-band.Proposed theoretically in [25,26] for the C-band, the U-shaped junction provides an efficiency similar to the SISCAP geometry because of its wide depletion region under reverse bias and large overlap between the change in the depletion region and the waveguide mode.The DC V π L of the junction is between 2.6 and 4.6 V•mm in the O-band, and the propagation loss is 12.5 dB/cm at 0 V bias.The junction is used to demonstrate a 24 Gb/s Si MZM with a 3 dB electro-optic (EO) bandwidth of 13 GHz and a 13 Gb/s microring modulator.These results compare favorably against state-of-the-art carrier accumulation MZMs [8], with the additional advantage that this junction has low loss.This modulation junction was fabricated on an 8" silicon-on-insulator (SOI) wafer as part of the multilayer silicon nitride (SiN)-on-Si platform in [27].

Overview
The U-shaped junction has a two-dimensional (2D) doping concentration profile in the waveguide cross-section, in which the P-type region extends into the N-type region near the vertical centre.The junction combines features of a lateral junction with a vertical junction.Figure 1 shows the doping concentration of the designed U-shaped junction for 0 V and -1 V bias computed using Sentaurus TCAD for the implant conditions and waveguide geometry to be discussed in this section.The blue regions are P-type and yellow/orange/red regions are Ntype.The red lines near the center of the rib waveguide mark the edges of the depletion region of the junction.For optimal efficiency, the P-type region, rather than the N-type, should be in the center of the rib since the refractive index change for a change in the density of holes is higher than that for electrons [1].
The junction works by carrier depletion.A high modulation efficiency requires the spatial change of the depletion region under an applied voltage to be large and maximally overlapping with the optical mode.Fig. 1(a) shows the depletion region covers only a small portion of the rib at a bias of 0 V, whereas in Fig. 1(b), with a bias of -1 V, the depletion region expands to cover most of the waveguide core.Thus, the U-shaped junction is efficient in producing an effective index change.

Junction Fabrication
The junction was defined in Si rib waveguides that were part of the photonic platform in [27].
The Si rib height was 150 nm and the slab height was 65 nm.The implantation steps to form the junction were specified and designed using Sentaurus Process.Table 1 summarizes the implantation steps.Boron and phosphorous were used for P-and N-dopants, respectively.The coordinates of the outer edges of each implantation window along the x-axis as labelled in Fig. 1 are given.To simplify the fabrication, the number of implantation steps was minimized, the implantation tilt angle was kept to 0° for all steps, and the implantation was designed to be alignment tolerant.A 0° tilt angle also allowed the PN diodes to be formed on curved waveguides, as in microring modulators.
To form the U-shaped junction, the Si waveguide was initially covered with a 10 nm thick layer of SiO 2 to reduce the channeling effect from the subsequent implantation steps.The key parameter in forming the junction is the implantation energies.In Table 1, the first step formed the P-type region in the middle of the waveguide using an intermediary energy of 20 keV.The second step with a high energy of 90 keV formed the N-type region at the bottom of the waveguide.The third step, with a low energy of 15 keV, formed the N-type region near the top of the waveguide.Conveniently, the two phosphorous implantation steps shared the same window to reduce the number of masks, and overall, the total number of implantation masks (i.e., four) is the same as required for a simple lateral PN diode.Together, these three implantation steps formed the U-shaped junction.Steps 4 and 5 formed the P++ and N++ regions for the contacts.Lastly, a rapid thermal anneal at 1030°C for 5 seconds activated the dopants.To improve the alignment tolerance of the implantation windows to the rib waveguide, we chose a Si rib width of 700 nm.The separation between the implantation windows and the edges of rib was 90 nm nominally.For the carrier profile cross-section in Fig. 1, the simulated optical loss is 14 dB/cm at 0 V bias, and the V π L is 0.29 V•cm at a bias of -1 V for λ = 1310 nm.TCAD and electromagnetic mode simulations consider up to ±60 nm misalignment of the low-doping P and N masks independently with respect to the silicon waveguide.The relevant P doping is step 1 of Table 1.The relevant N doping is steps 2 and 3 which share a mask.The worst case V π L and propagation loss are 0.35 V•cm and 18 dB/cm, respectively (occurring when the boron implantation window is shifted to the left by 60 nm, and the phosphorous implantation window is shifted to the right by 60 nm).The results show that the designed junction is tolerant to mask alignment errors.

DC Characteristics
The junctions were fabricated on 8" SOI as part of [27].Figure 2 shows the measured (using an LCR meter [Agilent E4980A]) and simulated diode capacitance as a function of the reverse bias.The measured phase-shifter was in a test structure with the same doping window positions as the bottom arm of the highest bandwidth MZM described in Section 3.3.In general, the measured capacitance and series resistance are in good agreement with the TCAD simulations.Compared to a lateral PN junction [2], the change in the capacitance of the Ushaped junction is greater for reverse bias between 0 V and 1 V, indicating the potential for higher efficiency.The simulated junction capacitance varies from 2.2 pF/mm to 0.34 pF/mm between 0 V and -1 V, and remains relatively constant beyond -1 V.However, at -1 V, the measured capacitance was 1.03 pF/mm, about 3 times higher than the designed value.The measured series resistance was 8.4 Ω•mm in agreement with the simulated value of 8.0 Ω•mm.The increased capacitance may have been caused by discrepancies between the simulated and fabricated doping profiles, and this affects the bandwidth of the MZM.

MZM travelling-wave electrode design
We designed traveling-wave electrodes in the single-drive push-pull geometry for MZMs with a target electro-optic (EO) bandwidth of 30 GHz [14,15].A 100 kΩ on-chip resistor (implemented in N-doped Si) and inductor (with a length of 6 mm and width of 2 µm wide implemented in the M1 layer) were added to the center DC line to isolate the DC bias from the RF signal [15].The phase-shifter length in the MZM was 2 mm.Fig. 3 shows the modulator cross-section with the dimensions of the designed electrodes.The grey regions are the doped Si rib waveguides of Fig. 1.M1 and M2 are aluminum layers.Metal vias connect the Si layer with the two metal layers.The RF drive signal is applied to the two outermost lines connected to the N++ regions, and the DC bias is applied at the center P++ region.This configuration (P++ in the center) reduces the contact resistance, since the contact resistance for the N++ doped region is less than that for the P++ doped region.
Travelling-wave electrode design should consider the velocity match between the RF and optical signals, the impedance match between the electrode characteristic impedance and termination resistor, and the RF loss [10][11][12].When the velocities and impedances are perfectly matched, the RF loss, α, at the EO S21 3dB frequency, f 3dB , is limited by α(f 3dB )•L=6.4dB, where L is the electrode length.The design of the electrodes was carried out using ANSYS HFSS.First, we computed the S parameters of single-drive push-pull electrodes.Then, the S parameters were simplified into an effective RLGC model.By loading the capacitance and resistance of the TCAD simulated U-shaped junction at -1 V bias (which were, respectively, 8.0 Ω•mm and 0.34 pF/mm), we calculated the propagation constant and characteristic impedance of the device [12].The widths and separations of the electrodes and metal vias were varied to converge to a design.The expected EO S21 of Si MZM is obtained by dividing the computed electrical S21 in decibels by 2, since the modulation of the optical power is proportional to the voltage.Fig. 4 shows the calculated RF refractive index, characteristic impedance, RF loss, and the EO S21 frequency response.The black curves show the results with the designed U-shaped junction capacitance and the red curves are the results with the measured capacitance at -1V.With the designed capacitance, the RF refractive index at 30 GHz matches well with the optical group index of 3.7; the characteristic impedance is around 50 Ω; and the RF loss is < 3.2 dB/mm at 30 GHz such that the total RF loss would be < 6.4 dB (for 2 mm long electrodes).Thus, the electrodes ideally should enable MZMs with bandwidths close to 30 GHz.However, the higher than expected junction capacitance compromises the bandwidth.The red curves show that with the measured junction capacitance the expected RF refractive index is > 4; the characteristic impedance is < 40 Ω, and the RF loss is < 3.2 dB/mm only at < 13 GHz.Fig. 4(d

M2
is about 28 GHz, but the bandwidth is reduced to 12 GHz with the measured junction capacitance at -1V.

MZM device
Figure 5 shows the optical micrograph of a fabricated Si MZM.It consisted of two 3 dB multimode interference (MMI) couplers separated by the waveguides with the U-shaped junctions.Single-mode Si channel waveguides routed light to and from the MMIs and adiabatic linear tapers were used to couple light between the routing waveguides and the 700nm wide, multimode U-shaped junction rib waveguides.A path length difference of 40 µm between the Mach-Zhender arms was implemented to enable measurements by varying the input wavelength.The measured device free spectral range (FSR) was 10 nm near a wavelength of 1310 nm.The RF drive signal was applied using a GS probe, and the travelling-wave electrodes were terminated off-chip using an SG probe with a 50Ω impedance.The DC bias was set by the DC pad.The device used inverse tapered edge couplers, and light was launched into and collected from the MZM using lensed fibers with a spot diameter of 2.5 μm.The IL of the test setup was normalized to the transmission on a straight waveguide on chip, and the device IL was found to be 2.7 dB.The phase-shifter loss was 12.5 dB/cm at 0 V for the highest bandwidth MZM to be described in Section 3.3.

DC characterization
To measure the DC tuning of the MZM, both the DC pad and the lower arm of the MZM were connected to ground, and a DC bias was applied to the top arm.The phase-shift was calculated from the shift of the transmission spectrum as a function of the applied bias.Fig. 6 shows the MZM transmission spectrum at different bias voltages for two devices with deliberately designed offsets in their doping window positions.Fig. 6(a) shows the result for the device with the highest bandwidth, and Fig. 6(b) shows the result for device with the highest efficiency.The extracted phase-shifts from the spectra are shown in Fig. 6(c).The device with the highest bandwidth had a boron implantation window designed with a +60 nm shift along the x-axis in Fig. 1 for the top phase shift arm.The device with the highest efficiency had a designed boron implantation window shifted +60 nm and phosphorous implantation window shifted -60 nm for the top phase shift arm.These results translate to a DC V π L of 0.26 V•cm at -0.5 V (red line) for the most efficient phase-shifter and a DC V π L of 0.46 V•cm at -0.5 V (black line) for the phase-shifter with the highest bandwidth.At a bias of -1 V, the average V π L of the two arms of the highest bandwidth MZM was about 0.61 V•cm; at a bias of -2 V, V π L was about 0.94 V•cm.At higher reverse biases, the efficiency diminished since the increase in the area of the depletion region reduced.

High-frequency characterization
We carried out S parameter and eye pattern measurements of the device with the highest bandwidth.The S parameters were measured using a vector network analyzer (VNA) (Agilent N5227A) and a 50 GHz photodetector (Finisar XPDV2320R).The RF cables and signal probe were de-embedded from the measurement (Agilent N4694-60001 E-cal kit, GGB Industries CS-8 substrate with SOLT calibration).Fig. 7(a) and (b) show the measured electrical S11 and EO S21 of the MZM at 0 V and -2 V bias.The wavelength was at the quadrature point of the MZM (-3dB optical transmission point) and input RF power was 0 dBm.The S11 was < -14 dB over a 30GHz bandwidth, showing the RF reflection was low.The ripples in the S11 were likely due to the external 50Ω termination, for which its RF probe was not calibrated.The EO S21 in Fig. 6(b) shows a 3dB bandwidth of 4 GHz at 0 V bias and extending to ~13 GHz at -2 V bias.The EO S21 is plotted as 10log(P opt /V in ), where P opt is the small-signal optical power modulation and V in is the small-signal input modulation voltage to the device, to compare with the eye diagrams, which were measured for optical (not electrical) powers.For this device, the EO bandwidth at -2 V bias was similar to that at -1V (curve not shown), which may be caused by a higher than expected capacitance in the phaseshifter of the top arm of the MZI at -2V bias (the result in Fig. 2 was for a test structure matching the bottom arm of the MZI). Figure 8 shows the measured eye diagrams of the device for 2 31 -1 pseudo-random binary sequence (PRBS) patterns.Drive signals from the pattern generator (SHF 78210D, 12104A) were amplified to 7.5 V pp (Microsemi OA4MVM3) and then attenuated.The drive signal amplitude was calibrated to be 2.88 V pp considering the attenuator and connector loss.Light from a tunable laser was amplified using an O-band semiconductor optical amplifier and bandpass filtered (passband bandwidth of 1 nm) prior to input to the MZM to overcome the IL of the device and setup.The eye patterns were captured using a sampling oscilloscope (Agilent 86100C, 86106B).For these measurements, the ER was sacrificed for the EO bandwidth as the junctions were biased at a strong reverse bias of -2.4 V to reduce the capacitance.Fig. 8(a) shows the eye pattern at 16 Gb/s with a RF voltage of 2.88 V pp .An extinction ratio (ER) of 2.6 dB was achieved with the input wavelength set at the MZM quadrature point.Figs.3(b) and (c) show the eye patterns at 20 and 24 Gb/s with the same bias conditions.The ERs were 2.4 dB and 2.2 dB at 20 Gb/s and 24 Gb/s, respectively.

Microring modulator
Because of the normal incident dopant implantation, we were also able to realize microring modulators using the U-shaped junction.The microring modulators, due to their compact sizes, did not require travelling-wave electrodes.Fig. 9(a) shows an optical micrograph of the ring modulator, which has a 65µm diameter and electrical pads in a GSG configuration.The P region was in the center of the ring.The DC tuning efficiency was measured using the shift of the microring transmission spectrum as a reverse bias is applied to the modulation junction.Figure 10(a) shows the EO S21 taken near the -2.55 dB transmission point using a VNA (Agilent N5232A) and a 38 GHz photoreceiver (Newport 1474-A).Input RF power was set to be 0dBm.The 3dB EO bandwidth was roughly 9.8 GHz at 0V bias, extending to 13.5 GHz at a bias of -1V.Fig. 10(b) shows a 13 Gb/s eye pattern for a PRBS 2 31 -1 pattern at a wavelength of 1310.58 nm captured using the sampling oscilloscope.The eye pattern is wide open.The ER was 10 dB at an IL of 2.5 dB (matching the EO S21 measurements) and was measured for a drive voltage of only 1.6V pp at 0V bias applied to a GSG probe.The measured bit rate was limited by the available pattern generator at the time of the experiment.

Discussion
Table 2 compares the results of this work with other phase-shifters and their implementations in MZMs.A figure of merit to compare the modulation sections is the loss-efficiency product, LEP, which is given by the product of the V π L (in V•cm) and waveguide loss (in dB/cm) as: (1) Compared to the other modulation phase-shifters, the U-shaped PN junction has the most optimal, i.e., the lowest, LEP between 3.25 and 5.75 V•dB.The LEP for the other phaseshifters range between 13 and 29.0 V•dB.The high efficiency of our demonstrated phaseshifter is the closest to the SISCAP carrier accumulation type.The major advantages are that the U-shaped junction has low optical losses similar to that of lateral PN diode phase-shifters and can be integrated in modulator geometries involving waveguide bends, such as microrings.Achieving a higher ER and dynamic modulation efficiency using the U-shaped junction would require reducing the reverse bias to use the high capacitance regime of the junction.However, a high capacitance leads to a high RF loss, as discussed in Section 3.1.To circumvent the issues of RF loss, as well as impedance and velocity matching, short phaseshifters that do not use travelling-wave electrodes can be used instead as in [8].
Table 3 compares our microring with other reported microring modulators.The refractive index change due to the Si free carrier plasma dispersion effect in the O-band is about 0.7 times that in the C-band [1], and the tuning efficiency of intracavity modulation is proportional to the cavity finesse [30].The cavity sizes and quality factors are variable in the demonstrations in Table 3.Therefore, to compare the relative modulation efficiency of the junctions, we define a figure of merit (FOM) that accounts for the differences in the wavelength-dependent Si plasma dispersion effect and the cavity finesse, where η is the resonance tuning efficiency in GHz/V, F is the finesse of the microring, and g = 0.7 for λ = 1550 nm and g = 1 for λ = 1310 nm.The FOM shows that the modulation efficiency of the U-shaped junction is superior to the vertical [19], interdigitated [24], and lateral PN junctions [31,32].Vertical and interdigitated PN junctions have similar efficiencies, while lateral PN junctions have lower efficiencies.In future design iterations, to improve the absolute tuning efficiency (i.e., η) of the microring with the U-shaped junction, the finesse of the microring should be increased by reducing the size of the ring.Simultaneously, the EO bandwidth can be boosted using rings with a broader optical linewidth.[33], which presents a similarly sized device.

Conclusions
In summary, we have demonstrated efficient and low-loss U-shaped PN junction phaseshifters for Si MZMs and microring modulators fabricated on 8" SOI as part of an integrated photonics platform.The fabrication of the junction is simple and does not require extra mask steps.The highest bandwidth Si MZM had a DC V π L of 0.46 V•cm at -0.5 V bias in the Oband and operated at up to 24 Gb/s.A DC V π L as low as 0.26 V•cm was measured.The microring modulator achieved a tuning efficiency of 40 pm•V -1 and operated at 13 Gb/s.Future improvements include re-designing the electrodes to suit the experimentally realized junction capacitance and resistance to increase the EO bandwidth and to boost the microring finesse to increase the tuning efficiency.The U-shaped PN junction has superior metrics compared to demonstrated modulator phase-shifters in Si.This work shows the promise of this type of PN junction for power efficient Si modulators.Such high modulation efficiency junctions can make possible very short MZMs that obviate the need for traveling-wave electrodes.

Fig. 1 .
Cross-sections of the doping concentration profiles computed using Sentaurus TCAD of the U-shaped PN junction under bias voltages of (a) 0V and (b) -1V.The waveguide height is 150 nm, the slab height is 65 nm, and the rib width is 700 nm.

Fig. 2 .
Fig. 2. Measured and simulated capacitance of the junction at different bias voltages.
Fig.4shows the calculated RF refractive index, characteristic impedance, RF loss, and the EO S21 frequency response.The black curves show the results with the designed U-shaped junction capacitance and the red curves are the results with the measured capacitance at -1V.With the designed capacitance, the RF refractive index at 30 GHz matches well with the optical group index of 3.7; the characteristic impedance is around 50 Ω; and the RF loss is < 3.2 dB/mm at 30 GHz such that the total RF loss would be < 6.4 dB (for 2 mm long electrodes).Thus, the electrodes ideally should enable MZMs with bandwidths close to 30 GHz.However, the higher than expected junction capacitance compromises the bandwidth.The red curves show that with the measured junction capacitance the expected RF refractive index is > 4; the characteristic impedance is < 40 Ω, and the RF loss is < 3.2 dB/mm only at < 13 GHz.Fig.4(d)shows that with the simulated capacitance, the expected EO 3dB bandwidth

Fig. 6 .
The MZM transmission at several bias voltages applied to the top arm for (a) the device with the highest bandwidth and (b) the device with the highest DC tuning efficiency.(c) The measured and simulated phase-shift as a function of the reverse bias voltage.

Fig. 7 .
Measured (a) S11 and (b) EO S21 of the MZM at a bias of 0V and -2V.The input wavelength was set to the quadrature point of the MZM.The S11 is < -14 dB over the 30 GHz bandwidth.The EO S21 3dB bandwidth is 4GHz at 0 V bias and extends to ~13 GHz at -2 V bias

Fig. 8 .
Measured eye diagrams using a PRBS 2 31 -1 pattern at (a) 16 Gb/s, (b) 20 Gb/s, and (c) 24 Gb/s.The input wavelength was at the modulator quadrature point near a wavelength of 1310 nm.The drive voltage was 2.88 Vpp.The device IL was 2.7 dB.The ER is 2.6 dB in (a), 2.4 dB in (b), and 2.2 dB in (c).

Fig. 9 (
b) shows the spectral tuning at a few bias voltages.The microring linewidth was about 50 pm (8.6 GHz) at 0V bias.The tuning efficiency was 40 pm•V -1 (7 GHz•V -1 ) between 0V and -0.5V near a wavelength of 1310 nm.(a) (b) Fig. 9. (a) Optical micrograph of the microring modulator.(b) Microring transmission at several bias voltages.(a) (b) Fig. 10.(a) The EO S21 of the microring for 0 V and -1 V bias taken at a transmission near -2.55 dB.The 3 dB bandwidth was about 9.8 GHz at 0V bias and extended to 13.5 GHz at -1V.(b) 13 Gb/s eye pattern of the microring modulator with a drive voltage of 1.6 Vpp at 0 V bias.