High-speed turbo-TCM-coded orthogonal frequency-division multiplexing ultra-wideband systems

One of the UWB proposals in the IEEE P802.15 WPAN project is to use a multiband orthogonal frequency-division multiplexing (OFDM) system and punctured convolutional codes for UWB channels supporting a data rate up to 480 Mbps. In this paper, we improve the proposed system using turbo TCM with QAM constellation for higher data rate transmission. We construct a punctured parity-concatenated trellis codes, in which a TCM code is used as the inner code and a simple parity-check code is employed as the outer code. The result shows that the system can o ﬀ er a much higher spectral e ﬃ ciency, for example, 1.2 Gbps, which is 2.5 times higher than the proposed system. We identify several essential requirements to achieve the high rate transmission, for example, frequency and time diversity and multilevel error protection. Results are conﬁrmed by density evolution.


INTRODUCTION
In recent years, ultra-wideband (UWB) communications has received great interest from both the academic community and industry. Using extremely wide transmission bandwidths, the UWB signal has the potential for improving the ability to accurately measure position location and range, immunity to significant fading, high multiple-access capability, extremely high data rate at short ranges, and easier material penetrations [1][2][3]. It is essential for a wireless system to deal with the existence of multiple propagation paths (multipath) exhibiting different delays, which are the result of objects in the environment causing multiple reflections on the way to the receiver. The large bandwidth of UWB waveforms significantly increases the ability of the receiver to resolve the different reflections in the channel. Two basic solutions for inter-symbol interference (ISI) caused by multipath channels are equalization and orthogonal frequency-division multiplexing (OFDM) [4].
OFDM is a promising solution for efficiently capturing multipath energy in highly dispersive UWB channels and delivering high data rate transmission. One of OFDM's successes is its adoption as the standard of choice in wireless personal area networks (WPAN) and wireless local area network (WLAN) systems (e.g., IEEE P802.15-03 [5], IEEE 802.11a, IEEE 802.11g, Hiper-LAN II). Convolutional encoded OFDM has been introduced as the proposed standard to combat flat fading experienced in each subcarrier [6]. The incoming information bits are channel coded prior to serialto-parallel conversion and carefully interleaved. This procedure splits the information to be transmitted over a large number of subcarriers. At the same time, it provides a link between bits transmitted on those separated subcarriers of the signal spectrum in such a way that information conveyed by faded subcarriers can be reconstructed through the coding link to the information conveyed by well-received subcarriers.
Trellis-coded modulation (TCM), proposed by Ungerboeck [7], is a well-established technique in digital communications for obtaining significant coding gains (3 ∼ 6 dB), while sacrificing neither data rate nor bandwidth. Recently compound codes have attracted much interest. Examples of such compound codes include turbo TCM (TTCM) [8][9][10], multilevel codes [11], and parity-concatenated codes [12,13]. Among the aforementioned compound codes, TTCM is an attractive scheme for higher data rate transmission, since it combines the impressive near Shannon limit errorcorrecting ability of turbo codes with the high spectral efficiency property of TCM codes. Different schemes using TTCM have been presented in the literature by several authors [8][9][10].
The basic idea in [8] is to map the encoded bits of a conventional turbo code (possibly after puncturing some of the parity bits to obtain a desired spectral efficiency) to a certain constellation. The decoding is performed by first calculating the log-likelihood ratios of the transmitted systematic  and parity bits, and then using conventional turbo decoding technology. The approach in [9] concatenates two simple trellis codes in parallel. The interleaver before the second trellis encoder operates on groups of information bits. A generalized decoding scheme used for decoding conventional binary turbo codes is used in this case. Benedetto et al. proposed TTCM in [10] where two component convolutional codes are used to produce parity-check bits, with the entire information block and its interleaved version as inputs. The outputs of the two component codes are punctured in such a way that only half of the systematic bits are outputted for the first component code and the other alternative half is outputted by the second component code. Then, the combination of systematic bits, together with the parity check bits from the component codes, is mapped onto a higher constellation. The MAP decoding algorithm is used in this scheme and achieves a performance better than two other schemes [10].
In this paper, we apply a TTCM encoder similar to that in [10] to examine the possible improvement for UWB/OFDM. We found a simple way to construct the encoder, equivalent to describing the turbo code as a simple repetition (i.e., the simplest parity-check code), an interleaver, and TCM, similar to the RA code structure [14]. Then, the bit MAP algorithm is applied in iterative decoding. The code performance is examined when applied to the OFDM systems in the UWB channel environments. Such a system can offer data rates of 640 Mbps via 16-QAM modulation and 1.2 Gbps via 64-QAM modulation. The code performance is confirmed by density evolution.
The paper is organized as follows. Section 2 presents the system description. Section 3 describes the coding and decoding scheme used in the UWB/OFDM system. Section 4 evaluates the code performance through density evolution. Numerical results are given in Section 5, followed by the conclusion section.

SIGNAL AND SYSTEM
In this section, we describe the coded OFDM (COFDM) signal and system and the UWB channel model.

OFDM system
The block diagram of the functions included in the COFDM system is presented in Figure 1. On the transmitter side, source information bits are first encoded and then mapped onto a higher-sized constellation, such as QPSK, 16-QAM, or 64-QAM. Then, the streams of mapped complex numbers are grouped to modulate subcarriers in OFDM frequency band. FFT and inverse FFT (IFFT) are used for a simple implementation [6]. IFFT is performed to construct the socalled "time-domain" OFDM symbols. After the IFFT at the transmitter, a certain length of trailing zeros is padded to avoid interblock interference (IBI). At the receiver side, the reverse-order operations are performed to recover the source information.
The FCC specifies that a system must occupy a minimum of 500 MHz bandwidth in order to be classified as a UWB system. The P802.15-03 project [5] defined a unique numbering system for all channels having a spacing of 528 MHz and lying within the band 3.1-10.6 GHz. According to [15], a 128-point FFT with cyclic prefix length of 60.6 nanoseconds outperforms a 64-point FFT with a prefix length of 54.9 nanoseconds by approximately 0.9 dB. Therefore, we focus on an OFDM system with a 128-point FFT and 528 MHz operating bandwidth. Subcarrier allocation can be found in [5].
We group 100 mapped 16-QAM or 64-QAM complex numbers to modulate 100 data carriers (or data tones) in an OFDM system with a 128-point FFT. Twelve of the subcarriers are dedicated to pilot signals in order to make coherent detection. Ten of the subcarriers are dedicated to guard tones for various purposes, such as relaxing the specifications on the transmitter and receiver filters. In a discrete-time implementation, 128 modulated subcarriers are mapped to the IFFT inputs 1 to 61 and 67 to 127. The rest of the inputs, 62 to 66 and 0, are all set to zero. After the IFFT operation, a length of D = 32 trailing zeros is appended to the IFFT output and a guard interval of length 5 is added at the end of the IFFT output to generate an output with the desired length of 165 samples.
Let C n denote the complex number vector corresponding to subcarrier n of ith OFDM symbol, which includes ith M × 1 information block s i M . Then all of the OFDM symbols s i M can be constructed using an IFFT through the expression below: A guard interval (T GI = 9.47 nanoseconds) ensures that only a single RF transmitter and RF receiver chain is needed for all channel environments and data rates and there is sufficient time for the transmitter and receiver to switch if used in multiband OFDM [15]. T FFT , T CP , and T GI make up the OFDM symbol period T sys , which is 312.5 nanoseconds in this case.

UWB channel model
Many UWB indoor propagation models have been proposed [16,17],The IEEE 802.15.3a Study Group selected the model in [17], properly parameterized for the best fit to the certain channel characteristics. There are two basic techniques for the UWB channel sounding-frequency-domain sounding technique and time-domain sounding technique. We use a frequency-domain autoregressive (AR) model [3] since it has far fewer parameters than the time-domain method and allows simple simulation of a UWB channel. As a result, the simulation model can be constructed and the simulation can be performed easily. The frequency response of a UWB channel at each point H( f n ) is modelled by an AR process: where H( f n , x) is the nth sample of the complex frequency response at location x, V ( f n ) is complex white noise, the complex constants b i are the parameters of the model, and p is the order of the model. Based on the frequency-domain measurements in the 4.3 GHz to 5.6 GHz frequency band, a second-order (p = 2) AR model is reported to be sufficient for characterization of the UWB indoor channel [3]. For a UWB model realization with the TR separation of LOS 10 m, the estimated complex constants b i could be The detailed parameters description can be found in [3]. The OFDM symbol blocks experience IBI when propagating through the UWB channels because the underlying channel's impulse response combines contributions from more than one transmitted block at the receiver. To account for IBI, OFDM systems rely on the so-called cyclic prefix (CP), which consists of redundant symbols replicated at the beginning of each transmitted block, or zero-padding (ZP), which are trailing zeros padded at the end of each transmitted block. To eliminate IBI, the redundant part of each block is chosen greater than the channel length and is discarded at the receiver in a fashion identical to that used in the overlap-save (OLS, for CP) or overlap-add (OLA, for ZP) method of block convolution. That means by inserting the redundant part in the form of CP or ZP, we are able to achieve IBI free reception. Furthermore, when it comes to equalization, such redundancy pays off. Each truncated block at the receiver end is FFT processed-an operation converting the frequency-selective channel into parallel flat-faded independent subchannels-each corresponding to a different subcarrier. Unless zero, flat fades are removed by dividing each subchannel's output with channel transfer function at the corresponding subcarrier. At the expense of bandwidth overexpansion, coded OFDM ameliorates performance losses incurred by channels having nulls on the transmitted subcarriers [18]. CP and ZP methods are equivalent and rely implicitly on the well-known OLS method as opposed to OLA. In the rest of this section, we will focus on IBI removal and postequalization of the Zero padded OFDM system over the UWB channel.
OFDM signal block propagation through UWB channels can be modeled as an FIR filter with the channel impulse In practice, we select M > D > L, where L is the channel order (i.e., h i = 0, for all i > L). Then, the expression of the ith received symbol block is given by where H is the P ×P lower triangular Toeplitz filtering matrix and H IBI is the P×P upper triangular Toeplitz filtering matrix as follows [19]:

EURASIP Journal on Wireless Communications and Networking
The IBI in this case is eliminated due to the all-zero We partition H into two parts: where H 0 represents its first M columns and H zp its last D columns. Then, the received P × 1 vector becomes since last D rows of F zp are all zeros. We then split the signal part in x i zp in (6) Padding M − D zeros in x i l and adding the resulting vector to x i u , we get where C M (h) is an M × M circulant matrix as follow: The noise will be slightly colored due to overlapping and addition (OLA) operation. Then, using FFT to perform demodulation, the received signal in the frequency domain is given by where

A simplified TTCM coding scheme
The TTCM was proposed by Benedetto et al. in [10]. Each of two component encoders has rate b/(b + 1) (where b is even), but only b/2 alternative systematic bits are selected to combine with the corresponding parity-check bit as the outputs for each constitutent encoder. The systematic bits for the second constitutent code are those systematic bits which are punctured in the first encoder. Two bit interleaves are involved in this TTCM encoder. The first interleaver permutes the bits selected by the first encoder and the second interleaves those bits punctured by the first encoder. For M-QAM, there are 2 1+b/2 levels in both I and Q channels, therefore achieving a throughput of b bps/Hz. One of the prototype of the 16-QAM TTCM is illustrated in Figure 2 A simple method can be used to describe the same code in Figure 3. This is equivalent to describing the turbo codes as a repeater (that is the simplest parity-check code), an interleaver, and one component code [14]. Two bit streams (u 1 and u 2 ) are provided at the input of the TCM encoderone is the original source information bit stream (u 1 ), and the other (u 2 ) is the interleaved version corresponding to the parity checks of the first one. TCM encoder has rate of 2/2, which combines only the original systematic bit (from u 1 stream) and the parity-check bit as the encoder outputs. Then, two consecutive clock cycle outputs (or two outputs after further interleaving) are mapped onto 16-QAM constellation-one for the in-phase component and the other for the quadrature component. If we make the interleaving size of the interleaver before the TCM encoder to be half of the information block size, the function of this concatenated structure is exactly the same as that of TTCM. Figure 4 illustrates the equivalence between TTCM and parity-concatenated TCM. Figure 4(a) is the 16-QAM block diagram of TTCM encoder with short block inputs u 1 = u 3 1 u 2 1 u 1 1 u 0 1 and u 2 = u 3 2 u 2 2 u 1 2 u 0 2 , whereas u 0 1 and u 0 2 are the LSBs and u 3 1 and u 3 2 are the MSBs. Assume after interleaving, two input sequences to the second constituent encoder are u 1 2 u 0 2 u 3 2 u 2 2 and u 2 1 u 0 1 u 1 1 u 3 1 , then 4 coded output sequences would be u 3 The similar coding results can be obtained through the encoder in Figure 4(b) with only a difference in the partial paritycheck bits. The merge from encoder of Figure 4(b) to that of Figure 4(c) is straight forward when we set the interleaver size and pattern as shown in Figure 4(c). Figure 5 describes the simplified encoder for 64-QAM modulation. There are 4 information streams (u 1 , u 2 , u 3 , and u 4 ) into the encoder. Among those four information streams, two streams (u 3 and u 4 ) are the interleaved versions of the original two source streams (u 1 and u 2 ), respectively. Again, two consecutive clock cycle outputs are mapped onto 64-QAM constellation via Gray mapping.
There are three advantages for such a modification. (a) We consider fewer interleavers: only one interleaver for the 16-QAM case (instead of 2) and 2 interleavers for 64-QAM case (instead of 4). (b) We save one constituent encoder. (c) It is easy to extend this to parity-concatenated codes. That is, we simply replace the repeater with a parity-check code.
When this coding scheme is applied to the OFDM system over the UWB channel, the coded bit stream is interleaved prior to modulation in order to provide robustness against burst errors. The bit interleaving operation is performed in two stages: symbol interleaving followed by OFDM tone interleaving. The symbol interleaver permutes the bits across OFDM symbols to exploit frequency diversity across subbands, while the tone interleaver permutes the bits across the data tones within an OFDM symbol to exploit frequency diversity across tones and provide robustness against narrowband interference.
We constrain our symbol interleaver to a regular block interleaver of size N P × number of encoder output bits, where N P is the input information packet length and the number of encoder output bits is 2 for 16-QAM and 4 for 64-QAM. The coded bits are read in columnwise and read out rowwise. The output of the symbol block interleaver is then passed through a tone block interleaver of size N OFDM × tone numbers in one OFDM symbol, where N OFDM is the OFDM symbol numbers for one packet and the tone number is 100 for the considered OFDM system. Still the coded bits are read in columnwise and read out rowwise.

Bit iterative MAP decoder
The MAP (maximum a posteriori probability) algorithm in iterative decoding calculates the logarithm of likelihood ratio (LLR), Λ(u b ), associated with each decoded bit u b at time k through (11) [20]: where P r {u b = i/observation}, i = 0, 1, is the a posteriori probability (APP) of the data bit u b . The APP of a decoded data bit u b can be derived from the joint probability λ i k (m) defined by where S k represents the encoder state at time k and y k is the received channel symbol. Thus, the APP of a decoded data bit u b is equal to From relations (11) and (13), the LLR Λ(u b ) associated with a decoded bit u b can be written as Finally the decoder can make a decision by comparing Λ(u b ) to a threshold equal to zero: The joint probability λ i k (S k ) can be rewritten using Bayes rule: in which we assume the information symbol sequence {u k } is made up of N independent input symbols u k with K input bits (i.e., u b , b = 1, . . . , K) in each u k and take into account that events after time k are not influenced by observations y k 1 and symbol u k if encoder state S k is known. For easy computation of the probability λ i k (S k ), probability functions α k (S k ), β k (S k ), and γ i (y k , S k−1 , S k ) are introduced as follows [21]: Yanxia Wang et al.

7
Then λ i k (S k ) can be simplified as The probabilities α k (S k ) and β k (S k ) can be recursively calculated from probability γ i (y k , S k−1 , S k ) through and γ i y k , S k−1 , S k can be determined from transition probabilities of the encoder trellis and the channel, which is given by (20) p(· | ·) is the channel transition probability, q(· | ·) is either 1 or 0 depending on whether the ith bit is associated with transition from S k−1 to S k or not, and π(· | ·) is the state transition probability that uses the extrinsic information of information u k . Using LLR Λ(u b ) definition ( Figure 10) and relations among λ i k , α k , β k , and γ i we obtain It was proven in [20] that the LLR Λ(u b ) associated with each decoded bit u b is the sum of the LLR of u b at the decoder input and of another information called extrinsic information generated by the decoder. Divsalar and Pollara [22] described an iterative decoding scheme for q parallel concatenated convolutional codes based on approximating the optimum bit decision rule by considering the combination of interleaver and the trellis encoder as a block encoder. The scheme is based on solving a set of nonlinear equations given by (q = 2 is used here to illustrate the concept [10,22]) L 1b = log u:ub=1 P y 1 | u j =b e uj L2j u:ub=0 P y 1 | u j =b e uj L2j , L 2b = log u:ub=1 P y 2 | u j =b e uj L1j u:ub=0 P y 2 | u j =b e uj L1j (22) for b = 1, 2, . . . , K representing b input bits per constituent encoder, where L 1 j are the extrinsic information and y q are the received observation vectors corresponding to the qth trellis code. The final decision is then based on L b = L 1b + L 2b , which passed through a hard limiter with zero threshold. The above set of nonlinear equations is derived from the optimum bit decision rule L b = log u:ub=1 P y 1 | u P y 2 | u u:ub=0 P y 1 | u P y 2 | u (23) using the following approximation: The nonlinear equations in (23) can be solved by using an iterative procedure where 0 corresponds to the all-zero symbol. The symbol MAP algorithm [21] can be used to calculate (26), as shown in Figure 6 [10]. Then the LLR of the bth bit within the symbol can be obtained by L b = log u:ub=1 e λ(u) u:ub=0 e λ(u) .

(27)
The symbol a priori probabilities needed in the symbol MAP algorithm, which is used in branch transition probability calculation, can be obtained by with the assumption that the extrinsic bit reliabilities coming from the other decoder are independent. In our case, we apply the turbo iterative MAP decoding scheme in [10,20,21], and make certain modifications to fit our concatenated encoder structure. For example, we only need one bit MAP decoder instead of two as in [10] for iterative decoding, since the outer parity-check encoders can be viewed as repeaters. So, the corresponding outer decoders only exchange extrinsic information between repeated bit streams. The decoder structure is depicted in Figure 7.
The bit MAP decoder computes the a posteriori probabilities P(u b | y, u) (y is the received channel symbol and u is the result from previous iteration), or equivalently the loglikelihood ratio Λ(u b ) = log(P(u b = 1 | y, u)/P(u b = 0 | y, u)). Then, the extrinsic information L e (u b ) out is extracted   from L e (u b ) out = Λ(u b ) − L c (u b ) − L e (u b ) in to avoid information being used repeatedly. It will be supplied to the paritycheck decoder. The outer parity-check decoder updates the L e (u b ) out into L e (u b ) in according to parity-check constraints between information bits and supplies it to the bit MAP decoder for the next iteration. L e (u b ) in is the extrinsic information, which is used as a priori probability for branch metric computation in MAP decoding process. L c (u b ) is the channel reliability for each u b .
Since half of the systematic bits from the inner TCM encoder are punctured, it seems that we can only get channel transition probability for the remaining half of the information bits and parity-check bits. However, the punctured information bits are the parity checks of those systematic bits at the encoder outputs except they are interleaved. So we can always find the channel transition probability for the punctured information bits through the unpunctured part. The extrinsic information value associated with π(·/·) in (20) is given as the logarithm format: If q(u b = 1/S k−1 , S k ) = 1, then otherwise π S k /S k−1 = 1 1 + e Le(ub) . (31)

DENSITY EVOLUTION FOR TTCM
Convergence analysis of iterative decoding algorithms is often used to predict code performance and to provide insight into the encoder structure. One of the methods-extrinsic information transfer (EXIT) method-has been widely used in particular [23][24][25]. The EXIT chart is a tool for studying the convergence of turbo decoders without simulating the whole decoding process. We use the density evolution method in [24] to confirm our simulation. We approximate the extrinsic information as a Gaussian variable whose mean is equal to half of the variance. In each iteration, we compute the average mean of the extrinsic information and then regenerate the extrinsic information as Yanxia Wang et al. 9 an independent Gaussian variable. Thus, the dependence between the extrinsic information bits is wiped out. This is the main difference between density evolution and simulation. Since TCM is typically irregular, density evolution using the all-zero sequence may be biased. So we need to consider both 0-bit and 1-bit as inputs which could bring a negative mean according to the definition of extrinsic information in (29). We examine the mean of extrinsic information using tens of thousands of randomly generated bit sequences and make it always positive regardless of bit sequences by weighting the sign of the bit. Such a mean can be easily traced by two decoding trajectories in the density evolution chart, that is, where overbar denotes the average. For UWB channels, we then average it over more than 2000 UWB channels.
Procedure of density evolution can be summarized as follows.
(1) Before the first iteration starts, all the extrinsic information is set to zero. (2) We divide each decoding process into two halvesone half-iteration for TCM followed by another half-iteration for parity-check codes. For each halfiteration, we calculate the updated extrinsic information through decoding. Using tens of thousands of simulation, we get the mean of the densities of those updated extrinsic information using (32). (3) Further, we assume the density to be Gaussian with the mean computed in (32) and the variance equal to twice the mean based on density symmetry condition [25]. Then, we regenerate the extrinsic information as an independent Gaussian variable for the next halfiteration. (4) During each half-iteration, SNR is estimated as half of the mean of extrinsic information. SNR, before and after each half-iteration, can then be tracked in the density evolution chart as in this paper.
The EXIT chart (see Figure 8)is plotted as a combination of two charts, one is for SNR1 in and SNR1 out and the second is for SNR2 in and SNR2 out . It shows input and output of extrinsic information (in terms of signal-to-noise ratio) for each decoder. Since the extrinsic information output of the first decoder is fed as the input for the second decoder, the combination of two charts easily demonstrates the convergent property of the code. For example, if two inputoutput curves are crossing, then the iterative decoder stops at the crossing point, which corresponds to a fixed nonzero error rate. Otherwise, the decoder produces extrinsic information with unbounded SNR, which corresponds to a zero error rate. The threshold is defined as the minimum value of signal-to-noise ratio that will generate two curves without a crossing point. Density evolution can be used to determine the threshold, which is the minimum SNR for the decoder to converge assuming infinite block length. In the density evolution chart, as long as the SNR is above the threshold, these two constituent transfer curves will never intersect, which  means convergence in the limiting case. A detailed description of density evolution and EXIT chart can be found in [23,25].
In Figure 8, we show density evolution for OFDM systems using 16-QAM on Gaussian and UWB fading channels. For Gaussian channels, we find the threshold is 2.6 dB and show the EXIT chart for E b /N 0 = 2.8 dB. On UWB channels, we find that if we take average over all 2000 channels, then EXIT chart shows the clear case of convergence (see curves with solid squares in Figure 8). However, if we run EXIT over each individual channel instance, then some channel instances require much larger SNRs to allow iterative decoding to converge to the correct codewords. For example, at E b /N o = 5.5 dB, about 2% of the channels are harder to converge (see curves with crosses in Figure 8). We call them "bad" channels. When E b /N o is small, the percentage of the worst channels increases significantly. For example, when E b /N o = 4.5 dB, about 20% of channels are bad. Good performance can only be achieved if the interleaver can fully randomize the extrinsic information over all channels. If the bits of a packet are interleaved over a number of channels containing significant amount of "bad" channels, then the performance will be much worse. This is the main reason that the packet error rate curve for UWB cannot drop sharply as those on AWGN channels. Figure 9 presents the density evolution analysis for OFDM system using 64-QAM on Gaussian and UWB fading channels. For Gaussian channels, we find the threshold is 3.7 dB and show the EXIT chart for E b /N o = 4.2 dB. For UWB channels, when we set E b /N o = 9.2 dB, about 2% of the channels are bad.

NUMERICAL RESULTS
The performance of the coding/decoding scheme is evaluated and applied to the OFDM systems for UWB channels. A similar simulation has been done over AWGN channels    Figure 10 shows the BER performance of the coded 16-QAM OFDM system and uncoded OFDM system in both UWB and AWGN channels as a function of E b /N 0 . Uncoded modulation scheme is QPSK in order to keep same system coding rate. For UWB channels, the line-of-sight (LOS) distance between the transmitter and receiver is 10 m. To measure BER at each point, we simulate up to 1.64 × 10 7 bits, which is 2000 packets ×41 OFDM symbols/packet ×100 QAM symbols/OFDM symbol ×2 bits/QAM symbol. The coded OFDM curve shows a big performance improvement over uncoded OFDM, especially on UWB channels. Furthermore, a BER of 8 × 10 −6 is obtained at E b /N 0 = 6.7 dB. Figure 11 describes the PER performance of the 640 Mbps coded OFDM system and uncoded case over UWB and AWGN channels. The low PER of 0.036 is obtained at E b /N 0 = 6.7 dB for coded OFDM over 10 m UWB channels.
The BER performance for 64-QAM coded OFDM system and 16-QAM uncoded OFDM system is illustrated in Figure 12. Again uncoded modulation scheme is lower than coded modulation scheme to keep the same system coding rate. There are 3.28 × 10 7 (2000 packets ×41 OFDM symbols/packet ×100 QAM symbols/OFDM symbol ×4 bits/QAM symbol) random bits simulated to measure the BER. The LOS of the UWB channel distance is 10 m. The simulation results indicate a BER of 2.3 × 10 −5 at E b /N 0 = 10.7 dB for 1.2 Gbps coded OFDM system over UWB channels. Figure 13 presents the PER performance for the same situation, reporting a low PER of 0.011 at E b /N 0 = 10.7 dB for 1.2 Gbps coded OFDM over UWB channels.

CONCLUSION
The proposed coding scheme utilizes a punctured parityconcatenated TCM encoder to function as a turbo TCM, which may have a big advantage in real-world implementation due to the savings of constituent encoder and Yanxia Wang et al.  interleavers. The scheme can be applied to the OFDM system over UWB channels for higher spectral efficiency. We identify several essential requirements to achieve high rate such as frequency and time diversity, and multilevel error protection. The simulation results show that we can deliver data rates from 640 Mbps up to 1.2 Gbps to current UWB systems. The results are confirmed by density evolution. In the future, we will extend the TTCM to parity concatenated TCM (i.e., replacing repeater to single parity-check code) for rate change, and perform a more accurate analysis on its BER upper bound and possible error floor.