Beamwidth-Enhanced Low-Profile Dual-Band Circular Polarized Patch Antenna for CNSS Applications

A low-profile dual-band circular polarized (CP) patch antenna with wide half-power beamwidths (HPBWs) is presented for CNSS applications. Simple stacked circular patches are used to achieve dual-band radiation. To enhance theHPBW for the two operation bands, a dual annular parasitic metal strip (D-APMS) combined with reduced ground plane (R-GP) is presented. A single-input feed network based on the coupled line transdirectional (CL-TRD) coupler is also proposed to provide two orthogonal modes at the two frequency bands simultaneously. Experimental results show that the 10dB impedance bandwidth is 32.7%.)e 3dB axial ratio (AR) bandwidths for the lower and upper bands are 4.1% and 6.5%, respectively. At 1.207GHz, the antenna has the HPBWof 123° and 103° in the xoz and yoz planes, separately. And the values are 127° and 113° at 1.561GHz.


Introduction
Nowadays, satellite navigation systems are intensively used in various fields, such as navigation, public safety, and surveillance. e compass navigation satellite system (CNSS), officially named as the BeiDou Navigation Satellite System, has achieved more and more attention due to the navigation and positioning services compatible with other systems [1]. To receive signals with stable capacity, most satellite navigation systems use circularly polarized (CP) antennas. ey have improved immunity to multipath distortion and polarization mismatch losses caused by Faraday rotation [2]. Among them, the CP microstrip antennas (CPMAs) have always been the research hotspot due to the advantages of low profile, light weight, and low cost. Besides, with the overall dimension of navigation system terminal getting smaller, compact CPMAs are highly demanded. Meanwhile, CPMAs with wide halfpower beamwidths (HPBWs) are urgently required to improve the coverage area and stabilize the received signal.
Generally, metal back cavity [3][4][5] or the similar structure of back cavities [6] is applied to enhance the HPBW of CP antennas. However, they suffer from high profile, and their complex in geometry may lead to fabrication difficulties. Recently, a parasitic ring is stacked on the radiation patch to effectively widen the HPBW to 140° [7]. Parasitic strips [8] are also proposed to achieve wide HPBW. Nevertheless, these technologies [3][4][5][6][7] are presented for single-band applications.
In [9], a dual-band CPMA with wide HPBW is reported. By extending the substrate beyond the ground plane, the HPBWs of more than 100°and 114°are obtained at the two center frequencies. But the impedance and AR bandwidths are narrow. In [10], a dual-band CP antenna with enhanced beamwidth is proposed. By using stacked cone patches and a dual-ring cavity, the HPBWs are 135°and 112°at the two center frequencies. In [11], a compact dual-band CP antenna with wide HPBWs is proposed by using four compact inverted-F monopoles, and cross dipoles combined with the cavity-backed reflectors are also presented [12]. However, high profile, complex in geometry, and high cost are generated by the structure. Currently, dual-band CPMAs used for GPS [13] or BeiDou [14] satellite navigation applications are reported. In [13], a modified metallic cavity is presented for wide axial ratio beamwidth. In [14], stacked patches with dual circular polarizations are proposed. But both of them ignore the enhancement of the HPBWs. erefore, it is essential to concentrate on improving the HPBW of a compact, low-profile dual-band CPMA.
In this paper, a compact dual-band CPMA resonates at CNSS B1 (1.561 GHz) and B2 (1.207 GHz) and is presented. To enhance the HPBW, a novel dual annular parasitic metal strip (D-APMS) combined with the reduced ground plane (R-GP) is presented. A compact single-input feed network based on the coupled line transdirectional (CL-TRD) coupler is also proposed to provide two orthogonal modes at the two frequency bands simultaneously. Detailed structures of the proposed antenna are presented in Section 2. In Section 3, the effects of the proposed D-APMS and R-GP are discussed and parametric studies are investigated. For demonstration, a prototype was fabricated and measured in Section 4. Comparisons are also presented between the design and some previous dual-band CP antennas, followed by a conclusion in Section 5. Figure 1 shows the structure of the proposed antenna. It consists of three layers of substrates (top, middle, and bottom), two radiation patches (stacked patches), the D-APMS, the R-GP, and the CL-TRD-based feed network. Each of the three substrates has a relative permittivity of 3, a loss tangent of 0.003, and a thickness of 1.5 mm.

Antenna Structure
As shown in Figure 1(a), the stacked patches are designed as circular patches. e small circular patch with a radius of R 1 , as the upper band main radiator, is printed on the upper surface of the top substrate. However, the big circular patch with a radius of R 4 , as the lower band main radiator, is printed on the upper surface of the middle substrate. To improve the HPBW at the two operation bands, a novel D-APMS is printed on the upper surface of the top substrate (same layer as the small circular patch). e D-APMS is formed by a small APMS with an inner radius of R 2 and width of d 5 and a large APMS with an inner radius of R 3 and width of d 6 . It is noted that the small and the large APMSs are divided into 4 sections by identical air gaps with the length of d 3 and d 4 , respectively. e small APMS is used for improving the HPBW of the upper band, and the large APMS is contributed to the lower band. Moreover, to further enhance the HPBW, an R-GP with a radius of R 5 is etched on the top of the bottom substrate, as shown in Figure 1 To provide two orthogonal modes on the stacked patches, a single-input feed network capable of simultaneously operating at the upper and lower bands is proposed, as shown in Figure 1(c). It consists of two CL-TRD couplers, a 90°phase shifter and a T-type power divider. e CL-TRD coupler, which is firstly introduced by Shie et al., can achieve tight coupling with weak coupled microstrip lines and allow decoupling the direct current path between the input and output ports [15]. Small size is also obtained compared with the branch-line coupler. In the design, a modified CL-TRD coupler [16] with improved power distribution and phase performance is applied to produce equal amplitude and consistent 90°phase shift. To suppress the mutual coupling between the upper and lower radiation patches, a 90°phase shifter is connected to the lower band CL-TRD coupler. Finally, a compact T-type power divider is used for connecting the two signal paths. As can be seen from Figure 1(c), the network provides four output ports (ports 2, 3, 4, and 5). e ports 2 and 3, with equal amplitude and 90°phase shift, are connected to the two short metal probes for feeding the lower band radiation patch, while the ports 4 and 5 are connected to the two long metal probes to feed the upper band radiation patch. us, two orthogonal modes on the two patches are excited, resulting in dual-band CP radiation waves. e input port (port 1) is connected to the coaxial cables. e modeling and simulation of the proposed antenna are performed with the 3D full-wave EM simulation software HFSS. Main dimension parameters of the antenna are listed in Table 1.

Effects of APMS and R-GP.
To investigate the effects of APMS and R-GP on the HPBW of the CPMA, a single-band circular patch antenna working at 1.561 GHz is simulated.
Here, four structures are compared, as shown in Figure 2. It starts from antenna 1, which is composed of a circular patch and a GP (the size of the GP is the same with that of the substrate). e antenna 2 is a circular patch antenna with an APMS etched on the same layer, while the size of the GP is also the same as that of the substrate. e antenna 3 is a circular patch antenna with an R-GP, and no APMS is used. e antenna 4 is a circular patch antenna with the combination of APMS and R-GP. It is noted that during the simulation, the dimensions of the substrate are fixed and the feed network is out of consideration. Moreover, the simulated results used for comparison are the optimal performances, including good impedance match (VSWR < 2), lowest AR, and widest HPBW. Figure 3 shows the simulated HPBWs of the antenna in the xoz plane. It is observed that the HPBWs for antennas 1 and 2 are the narrowest. When an R-GP is etched, the HPBW is widened to 91°. A widest HPBW of 101°is obtained when using antenna 4, which means that the combination of APMS and R-GP can enhance the HPBW effectively. e current distributions along the radiation patch and the APMS, from t � 0 to t � 3T/8, are shown in Figure 4. It is observed that the antenna with the APMS still maintains the CP radiation and the electric field flows in an anticlockwise direction, yielding a right-hand circularly polarized (RHCP) wave in the upper half-space. e loaded APMS around the circular radiation patch serves as 4 directors which can lead part of the electromagnetic energy to the sides of the antenna. erefore, the HPBW of the CP antenna is broadened.

Discussions of D-APMS.
It is demonstrated from Section 3.1 that the APMS contributes to the HPBW enhancement of the CPMA. However, one APMS is valid for one frequency band. In order to enhance the HPBW at two operation bands, the D-APMS is applied, which is composed of two APMSs. In this section, the location and structure of the two APMSs are discussed to obtain the optimal HPBW.
Firstly, the locations of the two APMSs are investigated. In order to prove that the D-APMS is better than APMS for dual-band HPBW enhancement, the dual-band CPMA with one APMS is also modeled and simulated. Figure 5 shows the different structures. In Figure 5 Since the APMS is served as a parasitic radiation which is fed by air coupling from the main radiation patch, the size of the small and large APMSs, as well as the spacing between them, is related to the energy coupling strength caused by the   International Journal of Antennas and Propagation 3 air gaps. During the simulation, to obtain the widest HPBW for each state, the sizes of the APMSs are optimized. Figure 6 shows the simulated HPBW in the xoz plane at the two center frequencies. e radiation patterns for the dual-band CPMA without the APMS are also plotted for comparison (gray dashed dotted line). It is observed that the HPBWs are    International Journal of Antennas and Propagation 97.2°and 92.1°for the lower and upper center frequencies, respectively. For the structure of Figure 5(a), the HPBWs at 1.207 GHz and 1.561 GHz are 98.8°and 108.4°, respectively, which indicate that the addition of small APMS contributes to the HPBW enhancement of the higher band and has less in uence on the HPBW of the lower band. Considering the structure of Figure 5(b), the values of the HPBWs are 111.6°a nd 87.3°, which state that the addition of large APMS contributes to the HPBW enhancement of the lower band and can reduce the HPBW of the higher band. us, for dual-band HPBW enhancement, D-APMS should be applied.
For the di erent locations of the D-APMS, the simulated results can also be found in Figure 6. It is observed that, at 1.207 GHz, the HPBW for location 1 is the narrowest. e HPBW for locations 2, 3, and 4 are nearly the same. And a widest HPBW of 132°is obtained for location 3. At 1.561 GHz, the structure of location 2 shows wider HPBW (130°) than the others, while the beamwidths for locations 1, 3, and 4 are closer. In a comprehensive consideration, the optimal structure is the structure of location 2, where the two APMSs are located on the upper surface of the top substrate.
Secondly, the locations of the gaps on the D-APMSs are considered, as shown in Figure 7. It starts from Gap 1, where the gaps on the small and large APMSs are in the directions of 0°, 90°, 180°, and 270°. For Gap 2, as shown in Figure 6(b), the gaps on the large APMS are the same with Gap 1, while the gaps on the small APMS are rotated by 45°. For Gap 3, the gaps on the small APMS remain unchanged and the gaps on the large APMS are rotated by 45°. Figure 8 shows the simulated results. It is observed that the in uence of the gaps on the small APMS is larger than that on the large APMS. At 1.207 GHz, the HPBWs for Gaps 1, 2, and 3 are 131°, 152°, and 108°, respectively. However, the values are 131°, 84°, and 85°at 1.561 GHz. us, the structure of Gap 1 is chosen.
Finally, the numbers of the gaps are discussed. Here, four states named as Strips 1, 2, 3, and 4 are investigated, as shown in Figure 9. For Strip 1, no gap is inserted to the APMS. For Strip 2, two gaps spacing 180°are added to the  Figure 10 shows the simulated HPBW at the two center frequencies. It is obvious that the antenna with Strip 3 shows the widest HPBW.

Parametric Study.
In order to investigate the influence of the APMSs and the R-GP, a parametric study is carried out using HFSS. Figure 11 Figure 11(b) shows the effect of the gap of the APMSs on the antenna HPBW. It is observed that as the gap of small APMS (d 3 ) increases from 6 to 8 mm, the HPBW at the 1.561 GHz is increased. However, when d 3 increases from 8 to 10 mm, the HPBW is decreased. During the change in d 3 , the HPBW at 1.207 GHz is stable. For the gap of the large APMS (d 4 ), peak HPBW is obtained at d 4 � 7 mm for 1.207 GHz. Figure 12 shows the effect of the gap between the two APMSs and the dimension of the R-GP. It is seen that when the gap between the two APMSs (d 7 ) increases from 4.0 to 8.0, the HPBW at 1.561 is decreased, while the peak value of 131°is obtained at 1.207 GHz when d 7 � 6.0 mm. For the dimensions of R-GP, the optimal value is 43 mm.

Measurement Results
To validate the proposed design, a prototype is fabricated. Figure 13 shows the photograph of the prototype. e overall size is 130 mm × 130 mm × 4.5 mm. |S 11 | of the fabricated antenna is measured by using an Agilent N5230A vector network  e far-field feature is measured in an anechoic chamber. Figure 14 shows the simulated and measured |S 11 | of the antenna. For |S 11 | < − 10 dB, the measured bandwidth is from 1.15 to 1.60 GHz. e differences between the simulated and measured |S 11 | may be caused by the fabrication errors. During the simulation, it is found that the gap error between the two patch layers may affect the performance of the antenna including |S 11 |. Besides, the value error of the shunt commercial capacitors is another reason. Figure 15 compares the measured and simulated gain and axial ratio (AR) at boresight. e measured minimum AR values of 1.3 and 0.6 are achieved at 1.21 and 1.53 GHz, respectively. For AR <3 dB, the measured bandwidths are 4.1% from 1.19 to 1.24 GHz for the lower band and 6.5% from 1.48 to 1.58 GHz for the upper band. In the frequency bands, the proposed antenna exhibits measured peak gains of − 1 dBic at 1.207 GHz and 4.2 dBic at 1.54 GHz. e gain at 1.561 GHz is − 0.6 dBic. It is observed that similar gains are obtained at the two resonated frequencies of CNSS. e reason for high gain at 1.54 GHz may be due to the HPBW reduction at the frequency point. From the analysis in Section 3, it is found that the gain is effectively increased with the decrease of the HPBW. Figure 16 shows the simulated and measured radiation patterns of the antenna at 1.207 and 1.561 GHz. It is observed that symmetrical radiation patterns are exhibited at the two center frequencies. At 1.207 GHz, the measured HPBWs in the xoz and yoz planes are 123°and 103°,  Table 2 compares our design with some previous dualband CP antennas. It is observed that the proposed antenna exhibits similar HPBW with other antennas [9][10][11][12] at the upper center frequency. However, at the lower center frequency, the HPBW of the proposed antenna is smaller than those in [10,11]. Although the antenna in [10] is better than the proposed structure, cavity has to be used, which increases design complexity and cost. In the item of impedance bandwidth and 3 dB AR bandwidth, the proposed antenna shows better performance than the antennas in [9,11,14]. However, compared with [10][11][12][13], the proposed antenna has smaller volume and lower profile. In summary, the proposed antenna features the characteristics of wider bandwidth and HPBWs with compact dimension and low profile.

Conclusion
In this study, a low-profile dual-band CPMA with wide HPBW is designed, fabricated, and measured. Wide HPBWs at the two bands are achieved with the proposed D-APMS and the R-GP. Moreover, with the employment of the CL-TRD-based feed network, wide impedance bandwidths are obtained. Among the published dual-band CP antennas with more than 100°HPBW for the navigation applications, the proposed CPMA has a wider bandwidth and a lower profile. erefore, the proposed structure could be a good candidate for the dual-band CNSS applications to improve the system angular coverage.

Data Availability
e data used to support the findings of this study are included within the article.

Conflicts of Interest
e authors declare that there are no conflicts of interest regarding the publication of this paper.