Beamwidth Enhancement of Microstrip Antennas Using Capacitive Via-Fence Loading

In this paper, a general approach to enhance the beamwidth of microstrip antennas is proposed for wide beam coverage in both E- and H-planes. A microstrip antenna loaded with two arrays of capacitive via fences is propounded and systematically studied. By introducing the vertical currents brought by the capacitive metalized vias, the half-power beamwidth (HPBW) is effectively broadened compared with the regular microstrip antennas. In addition, by utilizing the air medium with low loss, the microstrip antenna can be supported by the two arrays of via fences, maintaining a high radiation efficiency. To validate the proposed design, a prototype is fabricated and tested. The measurement results agree well with the simulated ones, with enhanced HPBWs of 100° and 90° in E- and H-planes, respectively. Compared with the existing 2.4-GHz antennas, the propounded antenna is with the advantages of wide beamwidth and high radiation efficiency, exhibiting the potential applications for space-limited mobile devices with wide coverage requirement.


I. INTRODUCTION
D UE TO the benefits of simple fabrication process and easy integration, microstrip antennas are widely utilized in the applications of communications [1], [2], [3], sensing [4], and RF harvesting [5]. By engineering various advanced structures, microstrip antennas have successfully achieved important properties for antennas, such as broad bandwidth [6], [7] and dimension miniaturization [8], [9]. However, for a typical microstrip antenna in a mobile device, the E-and H-planes' HPBWs is usually less than 80 • , limiting the applications of wide beam coverage. What's more, with the variation of the ground sizes of different mobile devices, the beamwidth also undergoes drastic reduction. Therefore, a feasible approach to enhance the beamwidth is required for microstrip antennas in mobile devices for wide beam coverage.
Different antenna design methods are developed to realize the wide beam coverage [10], [11], [12], [13], [14], [15], [16], [17], [18], and can be divided into five categories. The first one utilizes a thick substrate on the antennas [19], [20]. Two typical way of utilizing thick profile is by designing dielectric resonator antenna (DRA) and magnetoelectric dipole antenna. In [21], the authors utilize a dielectric resonator antenna (DRA) with an engraved groove and a comb-like metal wall for beamwidth enhancement in both E-and H-planes. In [22], the authors design a dual-polarized magnetoelectric dipole antenna with gain improvement at low elevation angle for a base station. In this way, by enhancing the radiation of low elevation, the beamwidth is effectively broadened [23], [24], [25]. In addition, by utilizing the thick profile, the bandwidth is enhanced at the same time. However, a thick profile will bring some effect to the antenna performance, such as an increase in antenna loss and a decrease in radiation efficiency. The second approach is to reshape the ground plane of the antenna, such as reducing the size of the ground plane [26], [27], [28] and introducing a 3-D ground plane under the patch [26], [29]. The beamwidth of the antenna can be broadened by making the floor edge current participate in the radiation. Accordingly, reducing the size of the ground leads to a front-to-back ratio less than 10 dB by introducing the strong back lobe. This approach is simple and convenient, but some application scenarios do not allow changing the size of the ground plane. The third method broadens the HPBW by reducing the size of the antenna [30], [31], [32]. By reducing the distance of two radiation apertures of the antenna, the broadside gain is decreased and the HPBW is broadened. This method can reduce the cost and size occupied by the antenna. However, reducing the size of the antenna will result in a reduction in bandwidth and efficiency. The fourth way to achieve beamwidth enhancement is by adopting parasitic structures around the main radiator [33], [34], [35], [36], [37]. The parasitic structure can effectively compensate for the radiation but makes the whole structure too large to use. In the fifth approach, surface waves can be excited via the introduction of a loaded high dielectric constant material to broaden the beamwidth [25]. However, utilizing a high dielectric constant medium usually lead to a narrow bandwidth which limits the application of antennas.
For traditional microstrip antennas, with the change in the ground size of smart devices, the beamwidth will undergo a drastic change and deterioration. Here, to solve this problem, we proposed a general method to achieve the beamwidth enhancement of microstrip antenna using capacitive viafence loading to compensate the ground effect. In this way, we can achieve the optimal beamwidth with a certain size of the ground plane. By applying the vertical current introduced by the metalized blind vias, the radiation pattern at low elevation angles can be compensated thus broadening the beamwidth. In this way, the beamwidth of the antenna is effectively enhanced without the occupation of large space and the decrease in bandwidth and efficiency. In addition, the overall antenna size is obviously reduced due to the capacitive loading effect introduced by blind vias. Furthermore, by employing the low loss air medium the total efficiency exceed 88% in the operating range of 2.39∼2.49 GHz to fulfill the requirement for high-speed and low-latency communication.

A. ANTENNA GEOMETRY AND STRUCTURE
As illustrated in Fig. 1, we first discuss the influence of ground size on the beamwidth of a normal microstrip antenna. We assume a microstrip antenna operated at 2.44 GHz, which is the center frequency of the Wireless Local Area Networks (WLAN) band, and λ 0 is the freespace wavelength at 2.44 GHz. W g1 is the ground size of the microstrip antenna. From Fig. 1, we can figure out that the ground size affects the HPBWs of both the E-and Hplanes of the microstrip antenna, and the E-plane HPBW is less than 60 • under most ground sizes and the beamwidth is too narrow to use. Our goal is to find a way to broaden

FIGURE 2. (a) Exploded view, (b) top view, (c) detailed view, and (d) cross-sectional view of the proposed antenna.
the beamwidth of both the E-and H-planes. Without the loss of generality, we choose a common and widely used ground size of 1.5 λ 0 to present our design. The geometric views of the propounded antenna are depicted in Fig. 2. The propounded antenna consists of a radiation patch, metalized blind via fences, two dielectric substrates, and a metal ground plane. An F4BM dielectric substrate with a permittivity of 2.65 and a loss tangent of 0.002 is used to manufacture the metalized patch. The blind vias are with diameter (D), period (p), and gap (g) to metallic ground. They are connected on both sides of the radiating apertures. There is no difference in size between these blind vias and the holes in the upper substrate. The printed ground is 2 mm thick on the back of the dielectric that supports the structure. Fig. 2(b) shows the metalized patch at the top and two substrates as squares with W and W g side lengths. In Fig. 2(c), the antenna is shown from a cross-section view. For simplicity, the bottommost substrate is made from the same F4BM material as the upper one. A 50-semi-grid cable is utilized to feed the antenna, with the inner and outer conductors soldered to the top patch and bottom metallic ground. Table 1 provides specific geometrical parameters of the proposed design. Here, Ansoft High Frequency Structure Simulator (HFSS) is used to optimize the antenna.
To clarify the evolution of the propounded approach,   compared with Case 1,the coverage area of Case 2 in the upper half plane has improved significantly. To be more specific, Fig. 6 depicts a comparison of the normalized E-plane radiation pattern of Case 1 and Case 2. It is clear that by applying the capacitive blind vias at the radiating aperture of a normal microstrip antenna, we not only double the E-plane HPBW from 52 • to 101 • but also broaden the Hplane HPBW from 74 • to 91 • . It is worth noticing that once applying the capacitive via-fence, the E-plane radiation pattern becomes unsymmetrical. To explain this phenomenon, Fig. 7 depicts the complex E-field magnitude distribution of the blind vias, the patch, and the ground plane. From Fig. 7(a), it is clear that the electric field magnitude on the two sides of the blind vias is not the same. The electric field strength is stronger on the blind vias which are closer to the feeding probe. In this way, the radiation pattern generated by the two sides of the blind vias is asymmetric. From Fig. 7(b), we can see that loading the blind vias will not affect the TM 10 mode of the patch antenna. Based on Huygen's principle, two magnetic currents lined up with a distance of d can approximately represent the radiation property of the antenna. Based on these two equivalent magnetic currents, the proposed antenna is able to achieve a broadside radiation pattern.

B. PRINCIPLE OF BEAMWIDTH ENHANCEMENT
Next, we discuss the principle of beamwidth enhancement of the proposed method. First, as we demonstrated above, the radiation of the patch antenna is equivalent to the radiation of two magnetic currents at the aperture. Therefore, the two in-phase magnetic currents construct a two-element array that can represent the radiation property of the patch. As we know, the radiation pattern of the array is equal to the product of the element factor and array factor. The array factor is given as 2 cos(π d cos θ), where d is the electrical length of the two magnetic currents and θ is the pitch angle between the z-axis and x-axis. We then assume the broadside radiation element factor as sin θ , so the radiation pattern of the proposed method can be written as 2 sin θ cos(π d cos θ). Therefore, we can figure that the distance between the two magnetic currentsaffects the radiation pattern and the beamwidth. In order to analyze the extent to which the distance between the two magnetic currents affects the beamwidth, here we plot the ideal radiation pattern in log format as Fig. 8. From Fig. 8, we can clearly see that when d varies from 0.2 λ 0 to 0.4 λ 0 , the radiation pattern and beamwidth only have a slide change. In contrast, when d switch from 0.4 λ 0 to 0.6 λ 0 , the radiation pattern show some obvious changes, such as the emergence of the sidelobe and dramatic deterioration of beamwidth Therefore, as illustrated in Fig. 8, we can conclude that from Case 1 to Case 2, the narrowing of current distance indeed has an effect on the enhancement of the beamwidth, but it is obviously not the dominant reason for that. As shown in Fig. 9(a), the current distributions on the proposed antenna are comprised of two parts: the horizontal current J h and the vertical current J v . The horizontal current is generated on the radiating patch, which forms a narrow beam radiation pattern with unidirectional radiation. Besides, the vertical current is generated on the blind vias, which forms a radiation pattern with ∞-shape. According to the superposition theorem of patterns, the radiation of the antenna at the low elevation angles is increased. Therefore, the beamwidth of the proposed microstrip antenna is broadened evidently. It is concluded that both reducing the distance of equivalent magnetic current and introducing the vertical current can broaden the beamwidth but the dominant factor for broadening beamwidth is the second one.

C. PARAMETER DISCUSSION
Next, some essential parameters of the propounded antenna are further discussed. Fig. 10 and Fig. 11 show the simulation E-and H-planes normalized radiation pattern with  various gaps (g). It can be seen that in either the E-and H-planes, the HPBW is getting larger when the gap gets smaller. As the g changes from 7 mm to 0.5 mm, the Eand H-planes' HPBWs are raised from 51 • to 118 • and 76 • to 96 • , respectively. The reason for this pattern is that the smaller the gap brings the larger the capacitance of the blind vias, which results in a larger vertical current appearing on the blind vias. As we have demonstrated, the generation of vertical current is the key to the beamwidth enhancement effect of the proposed method. What's more, the smaller the gap, the more asymmetric the electric field magnitude on the blind vias on the two sides. Thus reducing the gap will make the E-plane pattern asymmetric. According to the  conclusion in [38], the minor gap will bring a narrower bandwidth and the case of g = 1 mm is just enough to cover the bandwidth of 2.4-2.48 GHz. Therefore, in order to maintain the bandwidth and facilitate processing, we at last choose the g = 1 mm as the final result. Fig. 12 and Fig. 13 depicts the simulation E-and H-planes normalized pattern with various diameters (D). Adjusting diameter D from 1 mm to 2 mm results in the enhancement of the E-and H-planes' HPBWs from 89 • to 109 • and 89 • to 94 • , respectively. Since the greater diameter of blind vias can carry more vertical current on the surface, as a result, a wider beamwidth is produced for both E-and H-planes. Similar to the gap (g), greater D narrows the bandwidth of the proposed antenna, and thus for the same bandwidth consideration, we set D as 1.5mm. As shown in Fig. 14 and Fig. 15, changing the period (p) is only with a minor effect on the radiation of both the E-and H-planes.

III. PROTOTYPE AND MEASUREMENT RESULTS
Validation has been accomplished through the fabrication and testing of a prototype. The antenna is fed from a semi-grid cable as shown in Fig. 16. According to Fig. 17, the measured S 11 is in good agreement with the simulation results obtained with the N9917A vector network  analyzer. The simulation S 11 is lower than −10 dB from 2.39 GHz to 2.49 GHz and the measurement S 11 is lower than −10 dB from 2.37 GHz to 2.49 GHz. The minor variation between simulated and measured results is primarily a result of the processing error. In Fig. 18, the simulation and measurement total efficiency and realized gain are illustrated. The simulation realized gain is exceed 6.3 dBi in  the operating band. With the low-loss air cavity, the simulated total efficiency is above 91.5% in the operating range of 2.4∼2.48 GHz. The measured gain exceeds 6.1 dBi and the total efficiency exceeds 88.3% over the whole frequency range. Fabrication tolerance is mainly responsible for the minor variation between the simulated and measured results. On the basis of simulations and measurements, Fig. 19 illustrates the normalized radiation patterns of the proposed design at 2.44 GHz. As can be seen, the simulated and measured results are in good agreement. The propounded antenna exhibits broadside patterns with a wide beamwidth in both xoy plane (H-plane) and yoz plane (E-plane). The measured xoy plane (E-plane) HPBW reaches 100 • , whereas the measured yoz plane (H-plane) HPBW reaches 90 • . In addition, as shown in Fig. 19, the simulated and measured front-to-back ratio is greater than 20 dB, exhibiting low back lobes. The difference between the simulated and measured data is primarily caused by the measurement error of feeding cable. As a way of highlighting the ascendency and novelty of the proposed antenna, we draw a table based on antenna type, size, efficiency, and HPBWs of the Eand H-planes. Accordingly, the propounded antenna has a competitive ascendency over the antennas listed in Table 2, including its compact size and wide beamwidth of broadside radiation.
At the end of the paper, we want to add a method to solve the problem of the asymmetry of E-plane radiation pattern by applying the differential feed scheme. As shown in Fig. 20(a), the scheme uses two feeding points with equal amplitude and inverted phase. In this way, as shown in Fig. 20(b), the E-plane radiation pattern can restore to symmetric because the geometric structure becomes symmetric. However, the cost of eliminating the asymmetry of the radiation pattern is to add the differential feed network which will  make the structure complicated and lead to the reduction of antenna efficiency. Since the asymmetry of the radiation pattern is normal and does not affect the use of the antenna, we apply the single feed scheme in our manuscript eventually.

IV. CONCLUSION
This paper presents a method to enhance the beamwidth of the microstrip antennas. By utilizing the capacitive via-fence loading, the propounded antenna is with a stronger vertical current which can enhance the radiation of low elevation thus broadening the beamwidth. The approach is validated by fabricating, characterizing, and analyzing a prototype. Over the WLAN band, the measurement results are consistent with the simulation results, manifesting E-and H-planes' HPBWs of 100 • and 90 • . Thus, the proposed approach has the potential to pave a way for wide coverage and beam scanning.