A Reconfigurable Beamsteering Antenna Array at 28 GHz Using a Corporate-Fed 3-Bit Phase Shifter

A 3-bit beam steerable antenna array based on a microstrip line is designed and implemented at 28 GHz for millimetre-wave application. The antenna consists of an 8-element tightly stacked aperture and a switchable feeding network. The latter consists of a one-to-eight-way uniform microstrip corporate-fed power divider and is separated from the radiating element by a ground plane. To achieve coupled feeding, the ground plane has two rows of slots under each antenna element. A switching mechanism was implemented on the microstrip line to realize 3-bit periodic phase shifting for beamsteering. By sampling 8 feed points with a separation distance of $\lambda $ /8, a 3-bit phase shifter was constructed with phases of 0°, −45°, −90°, −135°, −180°, −225°, −270°, and −315°. Continuous beamsteering was achieved from −50° to50° by switching eight different sampling states, both in numerical calculation and in simulation. The design was verified by fabricating and testing three different prototype configurations at 28 GHz. The measured prototypes showed an excellent match with the simulation results. Notably, the maximum measured gain was found to be 13.44 dBi with a gain threshold of 10 dBi within the steering range (viz. 100°).


I. INTRODUCTION
M ILLIMETER-WAVE (mmWave) antennas have been widely explored for fifth-generation (5G) communication because of higher data rates and capacity [1]. The prime frequency band for 5G mmWave applications is 28 GHz, which has been allocated as an experimental band in different countries for the last few years [2]. However, due to the excessive path loss and atmospheric absorption of mmWave frequencies, a high gain antenna array is necessary for both transmitter and receiver systems to compensate for such losses [3]. In addition, wide spatial coverage, low power consumption, and low cost are critical metrics for mmWave applications due to the physical constraints and design challenges of mmWave antennas [4]. Concurrently, beamsteering is paramount to realizing the potential of this array due to the random orientation of the transmitter and receiver. As such, antenna elements must be fed by progressive phases to steer the beam to a specific orientation.
Traditional beamsteering techniques require external complex circuitry with multiple sources, resulting in systems that are bulky, lossy, and cost-prohibitive with severe temperature instability [5], [6]. To circumvent these complexities, different techniques have been implemented using feeding networks with delay lines [7], [8]. However, such techniques suffer from small scanning ranges and hardware complexity.
phase-shifting mechanism can be controlled by integrating tunable PIN diodes or digital switches. We note that these n-bit antenna arrays show superior performance over continuous phase shifting arrays because of their simplified biasing network, low losses, and low-cost [12].
A reconfigurable bit array can be divided into three categories. First, the majority of bit arrays constitute either reflect array [13], [14], [15], [16], [17], [18], [19], [20] or transmit array [21], [22], [23], [24]. These bit arrays can be reconfigured both in frequency and space under the illumination of radiation feed. The non-adjustable focal diameter ratio, feed blockages phenomenon, limited bandwidth [25], and complex reconfiguration mechanism of such antennas greatly restrict their use in space-related operations [26]. Second, reconfigurable bit antenna arrays based on travelling-wave or leaky-wave mechanism and microstrip have been gaining a lot of attention over the years for their low profile and tunability [26], [27], [28], [29], [30]. For instance, in [26], [27], a 2-bit phase shifter is used for the beamsteering using a microstrip-based leaky-wave antenna (LWA). It is worth mentioning that microstrip-based LWAs suffer from high dielectric losses [31]. Furthermore, since the antennas need an attenuator on the open end of the structure, the system suffers from additional power loss. In addition, the multiple reflections from each element drastically reduce the antenna efficiency [27]. Another drawback is that the steering angle of the travelling wave antenna varies with the operational frequency. Third, the feed-network-based bit antenna array loaded with phase shifters with each radiating element is considered one of the best candidates for beamsteering due to its flat structure [32], [33], [34], [35], [36], [37], [38]. In [32], a 3-bit corporate feed antenna array at 20 GHz was implemented with a moveable dielectric loading slab. The movement of the loaded slab changes the effective dielectric constant of the antenna and hence creates a progressive phase at each radiating element, leading to beamsteering in the desired direction. However, this antenna requires a complex hands-on operation. Further, the requirement for precise positioning of the slab makes the operation daunting and time-consuming. Therefore, in this paper, we present a 3-bit corporate-fed antenna array operating at 28 GHz that 1) eliminates the need for an attenuator, 2) has a steady steering angle within the operational frequency band, and 3) achieves easy operation for the desired beam direction compared to the existing 3-bit mechanism [32], [33]. A prototype was fabricated and tested showing high realized gain and a large steering range with a gain threshold of 10 dBi.
The paper is organized as follows. In Section II, we first introduce the corporate feed array geometry and theoretical knowledge of 3-bit phase quantization. We also show the design and simulation details of the 3-bit quantized phases as well as the beamsteering mechanism for the 8-element array. In Section III, we present the fabrication and testing of three prototypes. Section IV concludes the paper.   Fig. 1 and Fig. 2 show the 3-D exploded view and side view of the antenna array, respectively. The antenna consists of two tightly stacked substrate layers. The upper substrate is RT/duroid 5880 ( r = 2.2 and tanδ = 0.009), with a thickness of 0.79 mm. The lower substrate is RO4003 ( r = 3.55 and tanδ = 0.0027), with a thickness of 0.2 mm. The overall stack consists of three metal layers, as depicted in Fig. 2. The top layer of the upper substrate consists of the radiating element, which is a uniform patch array of 8 elements. The bottom layer of the upper substrate was etched completely. The ground plane is in the middle of the two substrates; hence, it is on the top layer of the bottom substrate. We note that the ground plane has symmetrically etched slots to achieve antenna feed coupling (see Fig. 1). The corporate-fed network is on the bottom layer of the lower substrate and consists of a one-to-eight-way uniform power divider. Each branch of the power divider consists of 8 extended branches from the two sides of the microstrip line and two side branches, as detailed in Fig. 3.

A. ANTENNA GEOMETRY
This side branch is connected to the extended edge of the microstrip line using a PIN diode. Thus, the eight extended lines, separated by λ g /8, represent the phases equivalent to a 3-bit phase shifter.  antenna. We note that eight switches are used to control the connection of the phase shifters. The 3-bit phase shifter topology is described in detail in the following section. Fig. 3 represents a typical 3-bit phase shifter topology using microstrip transmission lines. The 3-bit phase shifter can be realized by 8 extended edges on two sides of the transmission line. Four edges can be extended in one direction, and four edges can be extended from the opposite direction, forming eight extended edges on two sides of the transmission line. Here, edges a, b, c and d are forming a series (series-1) of the extended line, while a1, b1, c1, and d1 are forming another series (series-2) from the opposite side of the transmission line. The physical distance between the adjacent two edges is equivalent to λ g /8, which corresponds to the electrical phase difference of 45 • . Thus branch 'a' will be 0 • phase, branch 'b' will be −45 • phase, and 'c' and 'd' will be −90 • and −135 • phases, respectively. Because of the electric field's natural propagation, the phase difference between 'a' and 'a1' will be 180 • as both lines are extended from the same point of the transmission line. This can be easily identified that edges b1, c1, and d1 will have the phase of −225 • , −270 • and −315 • , respectively. This periodic phase can be mapped from 0 • to −360 • by using eight extended edges. These edges can be connected to the antenna by using 8 PIN diodes. The ON state of a PIN diode represents the corresponding phase of the antenna. Thus, eight binary switches can represent 3-bit phase shifter since a reconfigurable n-bit phase shifter can be built with 2 n switches. This phase shifter can play a vital role to steer the beam in a specific direction which will be detailed later in the subsequent section.

C. RECONFIGURABLE SINGLE ELEMENT DESIGN
A typical 3-bit phase shifter needs two antenna elements on either side of the transmission line, as shown in Fig. 3. Instead, we designed an aperture-feeding patch antenna, fed by a 3-bit phase shifter that only needs a single antenna element, as illustrated in Fig. 4. This feature distinguishes our presented antenna from a typical 3-bit antenna. The top substrate was chosen with a low dielectric constant and higher thickness to radiate power effectively in space, whereas the bottom substrate (see Fig. 4b) was chosen with a high dielectric constant and a thinner substrate to tightly bound the field in the transmission line. The radiation mechanism is dependent on the transmission line, where two side branches (side branch-1 and side branch-2) represent the excitation of the patch by coupling slots. The signal flows from the transmission line to the side branches when the corresponding switch is ON. Any switches from series-1 will excite side branch-1. Concurrently, any switches from the series-2 will feed to the side branch-2. As such, the energy in the side branch will be coupled to the corresponding coupling slot in the ground above the transmission line and excite the aperture. For a single element, whenever the switch is ON, the radiated field will be in broadside. The optimization of a single element depends on multiple variables that affect the coupling of the energy, such as the width of the coupling slot  (sw), length (cl), width (bw) of the side branch, and position of the coupling slot (sd). The optimized dimensions of the patch, slot and 3-bit phase shifter are shown in Table 1. Notably, the transmission lines employed in the design have a characteristic impedance of 50 .
As shown in Table 1, the distance between adjacent switches (point a to b or switch S1 to S2) is equivalent to 0.8 mm (λ g /8) for a −45 • phase difference. When switches S1 and S2, S1 and S3, and S1 and S4 are 0.8 mm, 1.6 mm, and 2.4 mm apart, the exact phase of −45 • , −90 • and −135 • can be achieved respectively. To demonstrate this, we tested the phases of the delay line by placing one port at point a and the second port at point b (λ g /8 =0.8 mm), c (λ g /4 =1.6 mm), and d (3λ g /8 =2.4 mm) and found the exact phases at 28 GHz which represents the ON state of the corresponding switches, as shown in Fig. 5. Because of the electric field's natural propagation, the phase difference between 'a' and 'a1' will be 180 • as both lines are extended from the same point of the transmission line. As a proof of concept, we investigated the near-field phase for both 'a' and 'a1,' which represent the ON state of switch S1 and S5. As shown in Fig. 6a, the observation point is 0.3λ 0 away to calculate the phase of the electric field. As shown in Fig. 6b, the phase difference is 180 • between two opposite switches S1 and S5, that are physically connected at the same point of the microstrip line. Similar results will be observed for 'b1,' 'c1,' and 'd1,' which are exactly 180 • phase apart from 'b,' 'c,' and 'd' respectively. Thus, our reported 3-bit phase shifter can provide the actual phase shift required for beam-steering.
Next, for the beam-steering array, each element is fed to a one-to-eight-way uniform power divider. However, the integration needs careful design optimization for reflection cancellation from the open end of the 3-bit phase shifter shown in Fig. 4b. This can be achieved using a 50 matched load at the open end of the 3-bit phase shifter at the expense of low gain and efficiency. The S 21 of the antenna with matched loads at 28 GHz is below −3.6 dB which indicates more than half of the input power is transmitted to port 2, as shown in Fig. 7a. Consequently, the realized gain becomes very low (< 1.7 dBi), as shown in Fig. 7b. Therefore, the matched load termination at the end of the 3-bit phase shifter is omitted from the design. Alternatively, a more versatile and effective solution is to use reflection cancellation stubs at the end of the 3-bit phase shifter. The simulation and validation of the antenna using reflection cancellation stubs are explained in the following section.

D. REFLECTION CANCELLATION STUBS
To improve the realized gain, maximum power should be coupled from the feeding line to the patch antenna. This is done using an open or short stub with the feeding line. The short stubs need vias and complex design and fabrication. Conversely, open stubs are convenient and simple to design and fabricate. The antenna has eight switching states, as shown in Fig. 3. As such, the length of the open stub (sl) will be different for each case since the feeding point is different for each switch along the transmission line. The optimized stub length (sl) is shown in Table 2 for the different switching conditions. We note that there are  Fig. 8a and Fig. 8b, respectively. Here, only the switches for phases with 0 • , −45 • , −90 • , and −135 • were shown, because switches from opposite sides lead to the same result. The gain pattern indicates that the radiation is in the broadside when the switch is ON.

E. BEAMSTEERING BY 3-BIT PHASE SHIFTER
As already mentioned, a beam-steering antenna can be built using a 3-bit phase shifter. To steer the beam in a specific direction θ , the required progressive phase distribution, s , for n number of elements can be written as follows Here, d is the distance between the antenna elements, k 0 is the wave number, and θ is the steering angle. We have eight antenna elements n = 8. First, the progressive phase s needs to be quantized and mapped from 0 • to −360 • in eight different states using the 3-bit phase shifter. The quantized phase distribution can be written according to the following principles for a 3-bit phase shifter. Finally, to steer the beam in a specific direction, every element of the antenna needs to be fed by quantized phases using (1) and (2). Here, each quantized phase distribution corresponds to the specific antenna configuration. We adopted a 3-bit phase shifter instead of a 2-bit phase shifter to minimize the phase quantization error. Because the maximum phase error of a 2-bit phase shifter is 45 • . This phase error is responsible for the mismatch of the main beam direction compared to the ideal scenario where no phase quantization was adopted. In addition, the expected side lobe level of the bit array is higher than the ideal case due to the phase quantization error.
To minimize this phase quantization error, we either need to increase the antenna elements significantly or increase the phase quantization (3-bit, 4-bit ).  in this quantized 3-bit phase shifter. It is worth mentioning that this error has very less effect to steer the beam in a specific direction. To analyze and prove this idea, we calculated the array factor for a uniform 8-element linear antenna array and fed them by quantized phases. The beam scanning was achieved by quantizing phases from −60 • to +60 • , as shown in Fig. 9. The next step is to integrate a one-to-eight-way power divider with an aperture-coupled antenna and steer the beam using the 3-bit phase shifting mechanism. An effective power splitting can be realized by a T-junction power divider. A one-to-eight-way power divider was designed to feed the 3-bit phase shifter. The width of the 50 transmission line is 0.44 mm and the width of the quarter-wave transformer is 0.24 mm.
The power divider was optimized to have equal power splitting with considerable copper and dielectric losses, hence S 21 was found as −9.3 dB to all the output ports, which is 0.3 dB lower than the ideal S 21 . The optimized spacing between the antenna elements is 0.56λ 0 . Fig. 10a and 10b show the feeding network integration with the 3-bit phase shifter and antenna with reflector, respectively. The realized gain in the broadside is 12.1 dBi, while the back lobe is considerably higher than 7 dBi. The feed network consists of right-angle bend, T-junction, and extended edges feeding the 3-bit phase shifter. These bends cause radiation from the corporate-feed network and hence a high back lobe of 7 dBi. To suppress this type of radiation, a metallic reflector was placed at λ 0 /4 distance underneath the corporate-fed network. Doing so, the broadside gain has increased by 2.3 dBi, while minimizing the back lobe level to −5 dBi. The simulated realized gain plots with and without the reflector are shown in Fig. 11. Likewise, reflector integration shows strong resonance near 28 GHz for the antenna array as shown in Fig. 12.
Using (1) and (2) the steering angles of the structure with reflector are calculated and simulated for different steering angles, as shown in Table 3. The steering angles were chosen arbitrarily at 0 • , ±13 • , ±26 • , ±39 • , ±43 • , and ±50 • . We found that the simulated beam agrees well with the calculations when the angle is near the broadside. When we approach end-fire radiation, the simulated beam becomes slightly less than the calculated beam. This discrepancy is expected and is due to the strong mutual coupling between the antenna elements when beam scanning approaches the end-fire direction. Fig. 13 shows the beam steering range from −50 • to +50 • at 28 GHz. Notably, the maximum beam error was found to be less than 5 • , as shown in Fig. 14. The maximum broadside gain is 14.42 dBi, while 10 dBi gain thresholds were observed from the steering range. The reflection coefficients of all steering angles are shown in Fig. 15, with an excellent 50 impedance matching at 28 GHz for all steering angles. It is worth mentioning that the beamsteering performance of this antenna is excellent at 27.5 GHz and 28.5 GHz as shown in Fig. 16 and 17. The realized gain for all scanning angles is maximum near 28 GHz and remains ≈ 10 dBi from 27.5 GHz to 28.5 GHz as shown in Fig. 18. Compared with the steering beam at 28 GHz, the beam of 27.5 GHz and 28.5 GHz are off by less than 5 • when the steering angle is greater than 30 • . As such our antenna exhibits a wide-angle coverage from 27.5 GHz to 28.5 GHz, making it a great candidate for 5G mmWave applications.

F. ARRAY ANALYSIS WITH ACTUAL SWITCH
To realize the practical switching for the 3-bit reconfigurable antenna array, a commercially available and widely used PIN diode (MACOM, GaAs based MA4GP907) [29], [39], [40] was used by incorporating its intrinsic electrical parameters in simulation. The diode was modelled as a 4.2 resistor  for the ON state, and a 0.025 pF capacitor for the OFF state according to the datasheet [41]. A DC biasing network is required to turn ON and OFF the diode by forward and reverse biasing it. A bias network consisting of a quarterwave transmission line and the radial stub was designed to block the RF signal as shown in Fig. 19a. The RF blocking is necessary to prevent antenna performance degradation due to DC biasing network. To do so, the input impedance of the biasing network must theoretically be infinite (viz. opencircuited) at 28 GHz. Fig. 19b shows the input impedance of the biasing network (viz. Z OC ) which is extremely high. Next, the DC biasing network needs to connect to the 3-bit phase shifter for actual circuit implementation. Fig. 20 shows the 3-bit phase shifter connected with the DC biasing network in two different positions of the side branch. When the DC biasing network is connected to the edge of the side branch, as shown in Fig. 20a, it drags a significant amount of current, causing the gain to decrease (see Fig. 21). Therefore, the geometrical position of the DC biasing network needs to be optimized to ensure that no current flows to the bias network. Fig. 20b depicts the DC biasing network optimized position at 0.19λ g away from the edge of the side branch. The comparison of realized gain for the different biasing positions is shown in Fig. 21. When the biasing network is on the edge of the side branch the realized gain drops significantly to 2.1 dB. Conversely, when the position is 0.19λ g away from the edge of the side branch, the realized gain perfectly matches the ideal condition with no biasing network.

FIGURE 19. a) DC bias network for 3-bit phase shifter b) Input Impedance of the biasing network representing an open circuit at 28 GHz.
To realize the 3-bit phase shift, eight diodes D1 to D8 (as switches S1 to S9) were incorporated with the phase shifter as shown in Fig. 22a. The capacitor C (0.1 μf) acts as a DC blocker to isolate the DC input from the RF input port. The yellow marked G1 to G8 between the diode and capacitor (C) acts as the negative terminal of the DC input. Voltage V1 is used to bias the diodes D1 to D4 and V2 is used to bias the diodes D5 to D8. Table 4 shows the different states of the DC voltage to turn ON a particular diode for the corresponding phases. Notably, only one diode will be ON during the operation of the phase shifter. The forward biasing current is set to 10 mA. When V1 is 0 V all diodes from D1 to D4 will be turned off. When V1 is +5 V and G1 is connected to the negative terminal of the DC source, D1 will turn ON. If G2 is connected, D2 will turn ON. Similarly, G3 will turn ON D3, and G4 will turn ON D4. Using this reconfiguration process, only the state of the negative terminal of the DC source (G1 to G4) is varied while the positive terminal (V1) is always kept at +5 V. The same procedure is adopted to turn ON D5 to D8 when V2 is set to +5 V.  Concurrently, to turn ON the diode D9 for connecting stub (sl) for the 2 nd group of Table 2, V3 is set to +5 V. Table 5 shows the different states of D9 to connect the stub (sl) with the 3-bit phase shifter. When D3 or D7 is turned ON the  stub (sl) must be connected for reflection cancellation hence D9 must turn ON simultaneously.
The equivalent circuit of the PIN diode is shown in Fig. 22b. When the diode is ON, it will act as a resistor (R). When it is OFF, it will act as a capacitor (C1). According to the datasheet of the diode, the impact of practical switches was evaluated by investigating the antenna performance. The latter greatly depends on the resistor of the diode during the ON state. Fig. 23a shows the realized gain of the antenna when the resistance is varied from the ideal case to 20 . The realized gain decreases significantly when the resistor value increases. The realized gain drops by 0.6 dB compared to the ideal case (R= 0 ) when a practical switch (R= 4.2 ) is employed in an equivalent circuit. Similarly, the effect of S 11 for different resistors was investigated as shown in Fig. 23b. The S 11 of the antenna shifts to the right side when the resistor increases from ideal to 20 , but still, it is lower than −15 dB at 28 GHz. Finally, the 3-bit array was re-simulated at 28 GHz using the equivalent circuit of the diode. Simulated beamsteering patterns are shown in Fig. 24. As seen, the overall realized gain drops by 0.6 to 0.7 dB without affecting the steering angle. As such, we presented a 3-bit reconfigurable array that is feasible in a real electronic environment that is broadly discussed from full-wave simulation.

III. FABRICATION AND ASSEMBLY
Three prototypes with different feed networks were fabricated, corresponding to steering angles of 0 • , +26 • , and +43 • . The fabrication was done using a standard milling machine. Fig. 25a, 25b, and 25c show the corporate-fed network at 0 • , +26 • , and +43 • , respectively. Fig. 25d shows the photograph of the top assembly of the 8-element patch array. Practical switches were not integrated for the connection for this proof-of-concept because of fabrication limitations. Instead, copper metallization was used to connect both sides of the switches. The substrates and a reflector were assembled by using four nylon screws and nuts. The spacing between the antenna and reflector was maintained by a nylon spacer. The antenna was measured by feeding it with a 2.92 mm connector.

A. MEASUREMENTS AND DISCUSSION
Figs. 26, 27, and 28 shows the simulated and measured S 11 for the three configurations (viz. 0 • , +26 • and +43 • ) depicted in Fig. 25  for all scanning angles due to fabrication inaccuracy and antenna assembly. We note that the substrate of these three antennas must be tightly stacked. Therefore, the small gap in the fabricated structure and the non-uniformity between the two substrates are the main causes of this minor mismatch between the simulation and measurement results. Fig. 29 shows the pattern measurement setup at Ka-band (26.5 GHz -40 GHz). The radiation measurement was done using a mmWave anechoic chamber. The radiation patterns at 0 • , +26 • , and +43 • at 28 GHz show measured realized gains of 13.42 dBi, 11.65 dBi, and 9.80 dBi, respectively. In addition, the measured half-power beamwidth (HPBW) is 10.1 • , 11.6 • and 13.7 • at 0 • , +26 • , and +43 • respectively. The simulated and measured results are in good agreement, as shown in Fig. 30. The radiation efficiency varies from 68% to 81% within the steering range.
When the steering angle increases, the mutual coupling between the antenna elements increases, implying a higher side lobe level (SLL), which is more noticeable at the +43 • steering angle. In particular, the expected side lobe level of the 3-bit array is higher than the ideal case due to the phase quantization error. Because, in the ideal case there is no  phase quantization, and it produces constructive interference at the desired angle. In a 3-bit array, the phase is quantized and rounded to the nearest quantized level (every 45 • ) that cannot maintain constructive interference, hence, resulting higher side level at the desired angle. This side lobe can be minimized using a tapered non-uniform excited corporate power divider.
The properties of the different antenna arrays operating at different frequencies are summarized in Table 6. Among them, our design demonstrates superior features in terms of scanning range, broadside gain, and gain threshold. For  Instance, our design shows higher realized gain and bandwidth compared to its close counterpart of leaky-wave series feed antenna [27]. In [27], a 2-bit 10 elements series feed travelling-wave antenna was realized with reflection cancellation stub in between the extended edges as these types of antenna suffer from the high mismatch. To steer the beam in a specific direction the phase delay of space and transmission line needs to be compensated for each antenna element on the leaky-wave array. In contrast, our design differs significantly from this leaky-wave antenna as it; 1) does not need to compensate for the phase delay as shown in (1), thus provides simple phase computation and operation; 2) has a corporate-fed network instead of series feed array 3) only one stub was integrated at the end of the transmission line instead of a bunch of stubs 4) does not need an attenuator 5) has higher realized gain and bandwidth.

IV. CONCLUSION
In this paper, we designed a 3-bit reconfigurable beam steering mechanism at 28 GHz for mmWave applications. The design consists of a tightly stacked aperture feeding antenna with eight reconfigurable quantized phases that can effectively steer the beam from +50 • to −50 • . The feasibility of a real electronic environment is simulated and broadly discussed to turn ON and OFF the PIN diodes for steering the beam within the scanning range (viz. 100 • ). Also, the 3-bit phase-shifting mechanism was realized using microstrip transmission lines to simplify the fabrication and assembly process. Our fabricated antenna arrays demonstrated a 13.44 dBi broadside gain with a 10 dBi gain threshold within the scanning range. Comparison with other reported mmWave antennae indicates the superiority of our design in terms of gain, spatial coverage, and bandwidth. As such our antenna is a leading candidate for mmWave and next-generation communications systems.