A High-Performance Isolated SEPIC Converter for Non-Electrolytic LED Lighting

This paper proposes a high-performance isolated single-ended primary inductor converter (SEPIC) that is integrated with a non-dissipative snubber for a light-emitting diode (LED) driver. The proposed LED driver employs the snubber circuit to minimize any switching overvoltage caused by parasitic leakage inductance of the high-frequency transformer in the isolated SEPIC converter. This paper also presents a detail theoretical analysis of the proposed LED driver including its operating modes and design calculation. The experimental results of the proposed LED driver at the rated power of 18 W shows that it has a high-power factor, low harmonics, employs non-electrolytic capacitors, low switching stress, high reliability and high efficiency.


I. INTRODUCTION
A light-emitting diode (LED) requires an LED driver to convert ac power to dc power. The LED driver is the likeliest to fail before the expected lifetime of the LED. Other causes of LED breakdown include housing, packaging and controller failures [1]. The risk of circuit failure in an LED driver may be caused by failure of electrolytic capacitor, switching overvoltage, and overcurrent.
A typical LED driver employs a line-frequency transformer in the circuit design to provide galvanic isolation. An indoor LED light with rated power of more than 15W and an outdoor LED light of any rated power will employ a linefrequency transformer to provide galvanic isolation in the driver [2].
In order to reduce the footprint of an LED driver, a highfrequency transformer is employed such as that in an isolated single-ended primary inductance converter (SEPIC) [3]- [5], flyback [6]- [8], push-pull [9], half-bridge [10], and The associate editor coordinating the review of this manuscript and approving it for publication was Amedeo Andreotti . full-bridge [11] converters. The flyback and SEPIC converters are simpler, more reliable and cost-effective compared to the other aforementioned converters because they use only one switch. However, the peak input current in flyback converter is high which increases the component's rating and cost. It also requires two different feedback controllers to solve the power factor and flicker issues [6]- [8]. On the other hand, SEPIC converter is able to achieve high power factor when its input inductor operates in discontinuous mode (DCM) operation which reduces the complexity and cost for a power factor correction (PFC) circuit. Fig. 1(a) shows the topology of an LED driver that is based on a snubberless isolated SEPIC. The switching frequency of isolated SEPIC is typically more than 50 kHz and highfrequency transformer T 1 is employed. However, the leakage inductance L lk in the high-frequency transformer T 1 causes overvoltage in the switch at every turn-off interval. Fig. 1(b) illustrates a 286 V overvoltage during switch turn-off in a practical 90 V ac-dc isolated SEPIC LED driver. The transient spike in switch voltage will increase switch voltage stress and switching loss in S 1 . Therefore, a snubber circuit is required to minimise the overvoltage to improve the reliability of the LED driver.
There are two types of overvoltage snubbers which are the passive and active snubbers. The latter type of snubber reduces the switching overvoltage by using an active circuit controller. The typical advantage of an active snubber compared to the passive snubber is zero-voltage switching (ZVS) or zero-current switching (ZCS). However, more switches and controllers are required which increases the size, cost and complexity of the overall system. It also compromises the reliability of the circuit.
The conventional passive snubber is known as a resistorcapacitor-diode (RCD) snubber [12]. It is a low-cost and straightforward solution to reduce the switching overvoltage. However, the snubber loss is not negligible, thus reducing the efficiency of the LED driver. To prevent losses in RCD snubber, an inductor capacitor diode (LCD) snubber has been used. The Domb-Redl-Sokal (DRS) snubber is a common non-dissipative LCD snubber for a flyback converter [13]. Nevertheless, the application of a DRS in an ac-dc isolated SEPIC converter for LED lighting is not yet available in the literature.
This paper proposes an LED driver that is based on a high-performance isolated SEPIC that utilizes a nondissipative DRS snubber [14]. The distinctive characteristics of the proposed LED driver are that it has a high-frequency transformer for isolation, minimised switch overvoltage, single-stage converter, high power factor and low harmonics. This paper also describes the topology of the proposed LED driver, its operating modes and design methodology. The simulation and experimental results validate that the proposed LED driver meets the strict requirements of a well-designed LED lighting system such as low harmonics, high power factor, high reliability and high efficiency.  Fig. 2 shows the proposed LED driver that is based on a nondissipative isolated SEPIC circuit. The dc input voltage V rec is the rectified average voltage of a single-phase full-bridge uncontrolled rectifier. The isolated SEPIC consists of an input inductor L 1 , a MOSFET switch S 1 , a storage capacitor C 1 , a high-frequency transformer T 1 , an output diode D o and an output capacitor C o . A non-dissipative DRS snubber that consists of snubber capacitor C sn , snubber inductor L sn and snubber diodes D 1 and D 2 are employed with the isolated SEPIC to improve circuit reliability. In practice, the parasitic leakage inductance of T 1 , L lk will cause overvoltage across switch S 1 at every switch turn-off. The DRS snubber is used to clamp the overvoltage across S 1 to an acceptable level.

II. CIRCUIT DESCRIPTION
To ensure a near-unity power factor, the current in inductor L 1 is in DCM so that the ac current waveform matches the ac source voltage waveform [15]. In the proposed LED driver, a feedback controller is not required for power factor correction. Therefore, the overall size and complexity of the LED driver is reduced. Transformer T 1 has a turns-ratio of 1:1 and it galvanically isolates the primary and secondary side of the proposed LED driver, providing protection to the circuit and users in the case of short-circuit faults. The magnetizing current of transformer i Lm is in continuous mode (CCM) with a low ripple to ensure that the ripple in output current I o is minimised. Capacitor C 1 attenuates the ripple of the rectified voltage. Output capacitor C o is a non-electrolytic capacitor, which reduces output flicker and prolongs the lifetime of the LED driver.  In Mode I, switch S 1 is turned on. Energy is transferred from the dc source and stored in inductor L 1 and the energy in capacitor C 1 is released to magnetizing inductor L m and leakage inductor L lk . Current in S 1 is the sum of current in L 1 and C 1 where,

III. OPERATING MODES
(1) No current flow in output diode D o as it is reverse biased in Mode I. Therefore, output capacitor C o supplies energy to the LED load. In the non-dissipative snubber, C sn resonates with L sn through the forward biased D 2 .
Mode II begins when switch S 1 is turned off at t = t 1 . The transformer magnetising inductor L m , leakage inductance L lk and capacitor C 1 discharge energy to snubber capacitor C sn since L lk resists the sudden change of current direction. As a result, the overvoltage in S 1 is minimized, thus increasing the reliability of the proposed LED driver. Energy in inductor L 1 is also discharged in C sn via diode D 1 . The duration for Mode II is very much shorter than the other operating modes due to the small value of C sn . In this mode, the energy in the primary side of transformer T 1 is transferred to the secondary side to charge output capacitor C o and to supply energy to the LEDs via diode D o . Meanwhile, D 2 is reverse-biased.
Mode III begins when C sn is fully-charged. At steady state, assuming that the effect of transformer leakage inductance is negligible, the peak voltage across S 1, V S 1 is expressed as, where N 1 N 2 = 1. From Fig. 4(a), i L1 decreases linearly to zero, where the current in L 1 is discontinuous. The dc input source and inductor L 1 transfer energy to capacitor C 1 and current flows in the primary winding of transformer T 1 . Magnetizing inductance L m discharges its energy into the primary winding of T 1 . Diode D o is forward biased, which provides a conducting path for the secondary winding of T 1 to transfer the energy to C o and LEDs. Mode IV begins when inductor current i L1 = 0. Since L 1 operates in DCM while L m operates in CCM, the value for inductor L m is much larger than L 1 . Therefore, energy from L m is transferred to the LEDs through output diode D o . When capacitor C 1 is fully charged, it blocks any current flow through it. The current in the magnetizing inductance is equal to the current in the primary winding 149896 VOLUME 9, 2021 of T 1 . Moreover, snubber diodes, D 1 and D 2 are reverse biased. Thus, no current flow in C sn and L sn .

IV. MODEL DESIGN A. INPUT INDUCTOR L 1
Input inductor L 1 has discontinuous mode of operation ensuring that the ac source current is in phase with the ac source voltage, given that switch S 1 is operated at a constant switching frequency and duty cycle. As a result, the power factor is high without requiring any PFC circuit or a feedback controller. From Fig. 4(a), the discontinuous operation begins when i L 1 becomes zero at t = t 3 . In the first approximation, inductor L 1 is designed to realize boundary conduction mode and the isolated SEPIC is assumed to be ideal. Therefore, the input power P in is equal to the output power P out and the equality is expressed as, where V rec is assumed to be the average output voltage of a single-phase full-wave uncontrolled rectifier, I L 1 is the average input inductor current, V o is the average output voltage and R e is the static equivalent resistance of the modelled LEDs. The average output voltage of the diode rectifier is expressed as, where V m is the maximum ac input voltage, while the average input inductor current I L 1 is determine from Fig. 4(a) as, where δ 1 and δ 2 are the durations for Modes I and III (the duration of Mode II is assumed to be negligible) and T is the switching period. The change in input inductor current i L 1 can be expressed as By substituting (6) into (5), the average input inductor current I L 1 is When substituting (4) and (7) into (3), the input and output power equality can be rewritten as Therefore, the input inductor L 1 is determined as, During the boundary conduction mode, δ 1 = (1 − δ 2 ). Therefore, for current i L1 to be discontinuous, the input inductor L 1 should be In DCM operation, δ 1 + δ 2 = 1.

B. MAGNETIZING INDUCTOR L m
The output current ripple, I o is proportional to the current ripple present in the magnetizing inductor of the transformer. Therefore, the transformer is designed to operate in continuous mode and low magnetizing inductor current ripple to ensure a smaller output current ripple and a smaller output capacitor C o . When switch S 1 is turned off, the magnetizing inductor current is expressed as On the secondary side, the diode current is expressed as, By substituting (12) into (11), the magnetizing inductor current i L m is rewritten as, In steady state, the average capacitor currents of C 1 and C o are zero. Therefore, the average magnetizing inductor current is Continuous current operation requires that the minimum magnetizing inductor current i L m,min > 0. At the boundary of CCM and DCM, i L m,min = 0. Therefore, When switch S 1 is turned on, Since the average voltage of capacitor C 1 is V rec , the voltage across magnetizing inductor when S l turned on is Therefore, the change in magnetizing inductor current is expressed as Substituting (14) and (18) into (15), In order for the magnetizing current to be CCM, the magnetizing inductance should be larger than the minimum magnetizing inductor from (14), therefore When switch S 1 turns on, the change in charge of C 1 is expressed as The minimum and maximum current of magnetizing inductor i L m,min and i L m,pk are expressed as By substituting (22) and (23) into (21), the change in charge of capacitor C 1 is The value of storage capacitor C 1 is expressed as the change in charge per unit in capacitor voltage, When substituting (24) into (25), The average output current I o = V o R e . Therefore, the storage capacitor C 1 is When switch S 1 turns off, Since the average voltage across the inductors are zero, therefore the average voltage of storage capacitor C 1 is, The ratio of change in voltage with average voltage of C 1 is expressed as, Substituting (29) and (30) into (27), the value of capacitor C 1 is,

D. OUTPUT CAPACITOR C o
When switch S 1 turns on, the voltage across the output capacitor C o is the same as the output voltage V o with ripple of double the line frequency (f l = 100Hz). The change of charge in capacitor C o can be expressed as where T l is double the line period with a value of 10ms. Rearranging (32), the expression for the output capacitor C o is,

E. NON-DISSIPATIVE SNUBBER DESIGN
The peak voltage of S 1 , v S 1,pk as shown in Fig. 4(b) is limited below the breakdown voltage of S 1 [13] v S 1,pk = v S 1 + i L lk where v S 1 is calculated as (2). Therefore, the value of the snubber capacitor C sn of the proposed converter is To prevent energy backflow into the source side during switch turn on, snubber capacitor C sn and inductor L sn shall not resonate so that the direction of current does not change and D 1 remains reverse-biased. Therefore, the maximum value of −v C sn,pk must be lower than rectified voltage V rec = 2V m π . This limitation can be used to derive the value for L sn as the following Note that the switch turn-on time T on is also constraint as [13], in order to ensure no back flow of current into the source.

V. SIMULATION AND EXPERIMENTAL VERIFICATION
A. SIMULATION RESULTS AT THE INPUT VOLTAGE OF 240V Fig. 7 presents the proposed LED driver that is modelled in Simulink R . The LED driver consists of an EMI filter, a frontend diode rectifier, a non-dissipative DRS snubber and an isolated SEPIC. The LED driver is connected to a singlephase 50Hz ac voltage supply, v s , that is varied between 90V and 240V to emulate a universal input. An inductorcapacitor L f C f filter is connected at the ac input of the diode rectifier to reduce harmonics in the ac current i s . The diode rectifier converts ac voltage to an unregulated dc voltage. The average output voltage of the diode rectifier, V rec provides input voltage to the primary side of the isolated SEPIC.   Table 1 presents the values of the circuit parameters employed in the proposed LED driver. An array of 9-by-4 3.3-V 1-W LEDs are connected to the output of the isolated SEPIC. The current in the LED array is designed to be 150 mA in each string at the rated power of 18W. No control of brightness has been considered in this study. Fig.8 presents the simulated ac input current and voltage waveforms of the proposed LED driver at input voltage 240V and the rated output power of 18W. The peak of ac input voltage is 339V while the peak of ac input current is 0.2A. An FFT analyser tool in Simulink R is used to measure the harmonics level in the input current. Fig. 9 shows that the harmonics levels in the simulated LED driver are less than the allowable limits in the IEC 61000-3-2 Class C standard.
The power factor of the proposed LED driver is measured by taking the ratio of load active power and source apparent power. The power factor obtained for the proposed LED driver at input voltage 240V is 0.92 which complies with the Energy Star standard that specifies the power factor of LED drivers to be higher than 0.9 for commercial LED light and 0.7 for residential LED light.      ac voltage supply is connected at the input of the LED driver. A digital signal processor is used to generate PWM signals between 0 and 3.3V, which are amplified between 0 and 15V by a gate driver. In order to maintain the output voltage of the LED driver at 30V when the input voltage varies between 90 and 240V, the duty cycle of the switch in the isolated SEPIC is varied between 0.07 and 0.21. Note that, the overall system of the prototype circuit is not optimized in terms of its size and arrangement. Table 2 shows the model and rating of components used in the LED driver prototype.   12 shows the measured waveforms of the ac current i s and ac voltage v s when the input voltage is 240V and the rated output power is 18W. The peak input current is 0.21A and the peak input voltage is 339V. The measured power factor is 0.924. Fig. 13 illustrates the measured power factor when the ac input voltage is varied from 90 to 240V. The results show that the proposed LED driver has power factor more than 0.9 within the universal input voltage range. The power factor complies with Energy Star guidance for commercial use.     The ac input current harmonics are measured using the analyzer function in digital phosphor oscilloscope Tektronix DPO4034B. It can be seen that harmonic contents from the proposed LED driver are lower than the set limits. Both the power factor and harmonics results show that the proposed LED driver has a good power quality. Fig. 15 shows the measured voltage waveform of switch S 1 , v S1 , when the input voltage 240V and the rated output power is 18W. Since the DRS snubber is employed, the    The flicker value of the 18W LED array is relatively high. However, it complies with the limit in CEC Title 24 JA8 standard. The output current ripple can be further reduced by increasing the value of output capacitor, C o . If lifetime of the LED driver is to be lengthen, then a film capacitor should be used because it has a longer lifetime limit than an electrolytic capacitor. However, the footprint of the output capacitor may increase. Therefore, there should be a trade-off in the LED driver design. Table 3 compares the simulation and experimental results of the proposed LED driver when the ac input voltage is 240V and the rated output power is 18W. The peak input current and power factor in the simulation and experimental circuits agree with each other, while certain odd harmonics in the simulation circuit is slightly lower than in the experimental prototype. Power factor and harmonics current levels meet the Energy Star guidelines and the IEC 61000-3-2 Class C standard, respectively. The maximum voltage and steady-state voltage during switch turn-off are higher in the experimental circuit, which could be caused by parasitic inductances that exist in the experimental circuit that was not modelled in simulation. The results from the simulation model and the experimental setup are similar.

C. SWITCHING OVERVOLTAGE REVIEW
Figs. 17 to 19 shows the switching voltage waveforms of the proposed LED driver with various ac input voltages. The measured steady state voltages of switch S 1 are 182, 214 and 286V when the ac input voltages are 90, 110, and 150V respectively. The maximum voltage in S 1 due to the over voltage ringing is 252, 288 and 368V as the ac voltage increases.    Figs. 20 and 21 present the switching voltage waveforms of the prototype without the non-dissipative DRS snubber. The ac input voltage is supplied at the same voltage of 110 and 150V. At steady-state, the voltage across S 1 is 224 and 298V as the ac input voltage increases. It can be seen from the figures that the maximum over voltage in the prototype is much higher without snubber. The maximum voltage in S 1 due to the over voltage ringing is 520 and 564V as the ac voltage increases. Table 4 summarizes the maximum voltage, overvoltage and the steady-stage voltage of the isolated SEPIC with and without the snubber circuit. The DRS snubber circuit provided the advantage of minimizing up to more than 100V of overvoltage in switch S 1 improving the reliability of the isolated SEPIC circuit.  The efficiency of the proposed LED driver is measured at different ac input voltage at the rated output power of 18W. The efficiency n is measured as, where P in = v s i s cos θ and P out = V o I o . The efficiency of the proposed LED driver ranges from 78% to 95%. The proposed circuit with the non-dissipative snubber has around 3% to 4% higher efficiency than the circuit without the snubber because the non-dissipative snubber reduces switching losses in S 1 by minimizing the switch overvoltage. The efficiency range of the proposed converter is comparable to the range that has been reported, which is typically between 79 to 96% for an isolated SEPIC based on various input and output voltages [4], [5], [16]- [19]. The efficiency of the proposed LED driver can be further improved if the PCB circuit is optimized in terms of its connection and arrangement. Table 5 compares the proposed LED driver with a range of isolated SEPIC converters that uses passive and active snubbers described in [4], [16]- [19]. The proposed LED driver is operated in universal input voltage range between 90V and 240V, achieves power factor of more than 0.9, peak efficiency of 95%, and uses a non-electrolytic output capacitor. There are very few published literatures that have tested the isolated SEPIC for LED application in universal input voltage. Moreover, the proposed non-dissipative DRS snubber reduces the switching overvoltage by more than 69% compared to isolated SEPIC without snubber as depicted in Table 4.
The LED driver in [16] has a high power factor that ranges from 0.98 to 0.99 and low output current ripple at only 18%. The type and value of output capacitor used has not been discussed. It utilizes a simple capacitor-diode (CD) snubber and is able to reduce 41% of the overvoltage during switch turn-off as compared to the isolated SEPIC converter without snubber application.
However, the overvoltage is double the steady state voltage, which required a large snubber capacitor. Therefore, the LED driver with the CD snubber is suitable for application with low VOLUME 9, 2021 input voltages. The peak efficiency is low at 82.3% when the output power is 32W.
The circuit proposed in [4] has presented a 30-V, 21-W LED driver with the power factor of 0.99. The odd harmonics complies with the IEC 61000-3-2 Class C standard, flicker is 25% which satisfies the limit set by California Energy Commission (CEC) Title 24 JA8 standard, non-electrolytic capacitors are used and the peak efficiency is 95%. Even though spikes are not present during switch turn-off in [4], the voltage stress in the switch is relatively high because the constant turn-off voltage is triple the input voltage. There is a trade-off between the power factor and switching voltage where a lower voltage stress in the switch would decrease the power factor due to smaller value of the storage and snubber capacitors.
The dc-dc converter of [17] uses coupled-inductor, auxiliary capacitor, auxiliary inductor and diode to reduce the switch overvoltage as well as remove the ripple in the input current. The overvoltage is 20% of the steady state voltage and the efficiency is improved by 1.3% compared to the isolated SEPIC with conventional RCD snubber. However, the couple inductor increases the footprint of the converter and a larger turns ratio will cause a higher conduction loss.
The proposed LED driver is also compared to the isolated SEPIC converters in [18] and [19], which are employed in high power applications of up to 2kW and 300W, respectively. In both applications, active snubbers were employed. The LED drivers comply to the Energy Star standard for power factor, have efficiency more than 90%, and harmonics levels that comply with IEC 6100-3-2 Class C standard. The overvoltage in both LED drivers is low at 7% and 8% of the steady state voltage, respectively. However, the usage of active snubbers increases the complexity and cost while reducing the reliability of the LED drivers. Moreover, the voltage stress in [18] are high at the main and snubber switches.

VI. CONCLUSION
This paper has presented a detailed description and circuit design of a high-performance isolated SEPIC converter with a non-dissipative snubber for an LED driver. The input inductor L 1 of the proposed LED driver has been designed to operate in DCM to achieve a high-power factor. In addition, the magnetising inductor L m of the transformer has been designed to operate in CCM to reduce ripple in the output current and voltage. With minimized output ripple voltage, the output capacitor C o can be a non-electrolytic-type, which results in prolonging the lifetime of the LED driver. The stepby-step design equations of the inductors and capacitors in the isolated SEPIC and snubber circuits have been explained in this paper. The simulation and experimental results showed that the proposed converter has power factor more than 0.9, the current harmonics complied with IEC-61000-3-2 Class C standard and an efficiency that ranges between 78% and 95%. A DRS snubber is used to minimize the switching overvoltage and increase the reliability of the isolated SEPIC. The experimental results revealed that overvoltage of the switch is minimized and the efficiency is improved by 3% to 4% at different input voltages in the proposed LED driver as compared to the isolated SEPIC without the snubber. He is currently lecturing on digital signal processing, robotics, and artificial intelligence at Universiti Tenaga Nasional, where he also performs research and consultancy in the same fields.
GIAMPAOLO BUTICCHI (Senior Member, IEEE) received the master's degree in electronic engineering and the Ph.D. degree in information technologies from the University of Parma, Italy, in 2009 and 2013, respectively.
In 2012, he was a Visiting Researcher at the University of Nottingham, U.K. From 2014 to 2017, he was a Postdoctoral Researcher and a Guest Professor at the University of Kiel, Germany. In 2017, he was appointed as an Associate Professor in electrical engineering at the University of Nottingham Ningbo China and the Head of the Power Electronics of the Nottingham Electrification Center. He was promoted to a Professor, in 2020. He is author/coauthor of more than 230 scientific papers. His research interests include power electronics for renewable energy systems, smart transformer fed micro-grids, and dc grids for the more electric aircraft. He is one of the advocates for DC distribution systems and multi-port power electronics onboard the future aircraft. Dr