A Low-RCS, High-GBP Fabry–Perot Antenna With Embedded Chessboard Polarization Conversion Metasurface

A Fabry–Perot (FP) antenna with high gain-bandwidth product (GBP) and wideband radar cross section (RCS) reduction is proposed using an embedded chessboard polarization conversion metasurface (CPCM). A polarization conversion metasurface (PCM) unit and its mirror unit are etched on both sides of a dielectric substrate with chessboard arrangement to form a CPCM. The in-band and out-of-band low-RCS features of the FP antenna are realized by destructive interference, and the high GBP is attributed to the FP resonant cavity. Moreover, an additional small CPCM designed for another PCM unit is embedded into the original CPCM to construct an embedded CPCM, which helps to improve in-band RCS reduction and has a positive effect on gain enhancement and 10-dB impedance bandwidth. The measured results show that the FP antenna with embedded CPCM has a 10-dB return-loss bandwidth of 8.48-12.21 GHz (36.1%), a GBP of 1338, a 3-dB gain bandwidth of 8.9-11.5 GHz (25.5%) and a maximum realized gain of 17.2 dBi. Meanwhile, the measured RCS reduction band covers 8–26 GHz (105.9%) with a peak RCS reduction of 21.2 dB at 9.4 GHz.


I. INTRODUCTION
Fabry-Perot antennas, also known as electromagnetic band gap (EBG) resonator antennas, Fabry-Perot resonator antennas, 2-D leaky-wave antennas, resonant cavity antennas and partially reflecting surface (PRS) antennas, have recently attracted considerable attention due to their highly directive radiation pattern and simple design [1]- [5]. The FP antenna was pioneered in the work of Von Trentini in 1956 [1]. Subsequently, a series of articles on FP antennas were published which contributed to the improvement of the FP antenna The associate editor coordinating the review of this manuscript and approving it for publication was Santi C. Pavone . performance in different aspects, including the optimization analysis [2]- [4], theoretical interpretation [5]- [8], and optimization design [9]- [12]. Many properties can be achieved with FP antennas, such as low profile [9], multi-band [10], circular polarization [11], RCS reduction [12], reconfigurable [13]- [15] and wideband [16] operation. With the rapid development in science and technology applications, wideband and RCS reduction properties of antennas play more important roles in many communications and sensor systems for some stealth platforms.
Nowadays, many achievements have been made in wideband FP antennas. The 10-dB return-loss bandwidth, gain enhancement, and gain-bandwidth product (GBP) of VOLUME 8, 2020 This work is licensed under a Creative Commons Attribution 4.0 License. For more information, see https://creativecommons.org/licenses/by/4.0/ FP antennas have been greatly improved. Frequency-selective surface (FSS)-based PRSs are usually employed to achieve wideband characteristics for FP antennas [15]- [24].
In [15]- [19], single-layer FSS-based PRSs are used to achieve relatively large GBPs for FP antennas. FP antennas with single-layer FSS-based PRSs have merits of simple assembly and low profile, but their GBPs (less than 672) should be further improved. In [20]- [24], multi-layer FSS-based PRSs are presented to improve the gain enhancement for FP antennas. Most of them obtain high gains, but the GBPs are not raised enough. Meanwhile, it is more difficult to extend the 10-dB return-loss bandwidth and 3-dB gain bandwidth of FP antennas by using multi-layer FSS-based PRSs due to increased design complexity. All dielectric-based PRSs exhibit excellent capabilities in bandwidth expansion and wideband gain enhancement for FP antennas [25]- [40]. In [25]- [27], single-layer dielectric PRSs are employed to realize wide 3-dB gain bandwidths of FP antennas. Generally, single-layer dielectric PRS FP antennas achieve high GBPs by improving the radiation performance of the source antenna, which increases the design complexity of the source antenna. In [28]- [39], multi-layer dielectric PRSs are utilized to acquire high GBPs of FP antennas. Several types of multi-layer dielectric PRSs are adopted to ameliorate the radiation features of FP antennas, such as untruncated multi-layer dielectric PRSs [28], [29], truncated multi-layer dielectric PRSs [30]- [36], and transverse permittivity gradient (TPG) multi-layer dielectric PRSs [37]- [39]. Most of the multi-layer dielectric PRSs exhibit a strong capability in improving the GBP, but they may introduce higher longitudinal profiles compared with single-layer dielectric PRSs. Also, in [40], a near-field correcting structure (NFCS) all-dielectric PRS is used to optimize the wideband gain enhancement, which can locally modulate the phase of each correcting region of the antenna. Although the NFCS can effectively improve the GBP of the FP antenna, it also causes a large longitudinal profile.
Recently, metasurfaces have attracted tremendous attention in RCS reduction and bandwidth expansion for FP antennas due to their superior capability in electromagnetic wave modulation. In [12], [14], [41]- [44], low-RCS FP antennas are designed by constructing resistor-loaded absorbing surfaces (ASs) and PRSs. Although an FP antenna that is constructed by AS can obtain a wider RCS reduction band, its GBP is relatively low because some radiation energy of the antenna is absorbed by the AS. Chessboard arranged metasurfaces (CAMs) and polarization conversion metasurfaces (PCMs) are also exploited to reduce the RCS for FP antennas [45]- [48]. Both approaches depend on two FSS units with a reflection phase difference of 180 • ±30 • and etched on a single-layer dielectric substrate in a chessboard arrangement. From the mechanism of destructive interference, these methods have the potential to reduce the RCS over a wide band and improve gain enhancement. However, it can be observed in [45]- [48] that FP antennas designed with CAMs or PCMs fail to produce high GBPs. Therefore, it is necessary to find a new approach to realize simultaneously wideband RCS reduction and high GBP for FP antennas.
In this paper, an embedded CPCM is proposed to achieve wideband low RCS and high GBP for an FP antenna. First, a CPCM is designed and investigated. The in-band and out-of-band low-RCS features of the FP antenna are determined by destructive interference, and the high GBP is attributed to the FP resonant cavity. Secondly, an additional small CPCM replaces the central tile of the CPCM to construct an embedded CPCM. The embedded CPCM can effectively improve in-band RCS reduction. Also, it has a positive effect on the gain enhancement and 10-dB return-loss bandwidth of the FP antenna. Measured results confirm the validity of the design which is presented in the following sections.

II. ANALYSIS OF POLARIZATION CONVERSION METASURFACE A. DESIGN OF POLARIZATION CONVERSION METASURFACE
According to the theory of FP antennas, when the FP resonant condition (1) is satisfied, the cavity resonance between the CPCM and the ground of the source antenna can be excited, and the directivity of the FP antenna can be calculated by using (2) [1].
In (1) and (2), ϕ CPCM and ϕ GND represent the reflection phases of the CPCM and ground, respectively; h c is the height of the cavity, and c is the speed of light. D and D p denote the directivities of the FP antenna and source antenna, and R is the reflection amplitude of the CPCM. In order to design a high-gain FP antenna with a large 3-dB gain bandwidth, the reflection amplitude of the CPCM should be as large as possible, and a positive reflection phase gradient is required in the operating band [18].
A PCM unit and its mirror unit will be designed with chessboard layout to form the CPCM, which achieves both wideband FP resonance and RCS reduction for a wideband FP antenna. The positive reflection phase gradient band of the unit needs to match the operating band of the primary antenna in order to excite the FP resonance. The in-band RCS reduction of the FP antenna depends on the mutual cancellation of co-polarized reflected waves and co-polarized retransmitted waves of the PCM units, while the out-of-band RCS reduction is attributed to the polarization conversion characteristics of the PCM units. Also, the RCS reduction is related to the design of the CPCM. The number of CPCM tiles, the number of units that make up the tiles, and the arrangement method have an effect on the RCS reduction. Therefore, the design of the PCM unit structure is determined after considering various factors. For this purpose, a PCM 1 unit and its mirror unit are designed. The PCM 1 unit is composed of a square metal patch and a dumbbell structure (Fig. 1). These units are etched on both sides of a F4BM265 (ε r1 = 2.65, tan δ = 0.0007) dielectric substrate. As the incident wave irradiates the PCM 1 along the +z direction, the square metal patch at the bottom of the PCM 1 exhibits the characteristic of a wide band positive reflection phase gradient, which improves the wideband gain enhancement for the antenna through the FP cavity resonance. As the incident wave irradiates the PCM 1 along the -z direction, the top structure can make the polarization conversion of the reflected wave in a wide band. The phase difference between the reflected waves of the PCM 1 unit and its mirror unit is 180 • . The FP antenna to be designed uses destructive interference to achieve the wideband RCS reduction characteristic. As the phase difference between two waves traveling in the same direction is close to 180 • ±30 • , destructive interference will occur [45]. Fig. 2 shows the operating mechanism of the destructive interference on the CPCM. After the PCM 1 and its mirror unit are arranged on a chessboard to form a CPCM, as the two incident waves irradiate the different units of the CPCM, the reflected waves will produce destructive interference because the phase difference of the two reflected waves is 180 • ; then the energy of the two reflected waves will be eliminated, thus achieving wideband RCS reduction for the FP antenna. All dimensions are optimized as follows: p = 4.8 mm, r 1 = 2.2 mm, w 0 = 0.5 mm, w 1 = 1.0 mm, g = 0.1 mm and h 1 = 3 mm.
The properties of the PCM 1 are analyzed by the equivalent circuit method (ECM) and CST simulations. Based on equivalent circuit theory, the PCM 1 exhibits dual diagonal symmetry that can be described by a four-port equivalent circuit [49]. The S-parameters are related to the four-port network; as a diagonal excitation is applied, the four-port network must be simplified to parallel admittances. When x or y polarized waves irradiate the PCM 1 along the -z direction, they will generate both field components of u and v incident polarizations. The equivalent circuit of the PCM 1 is depicted  in Fig. 3. The bottom structure and dielectric of the PCM 1 are equivalent to an LC series circuit and a transmission line, respectively. The top structure is equivalent to connecting a quadripole matrix T c , which transforms the polarization direction of the incident wave. The expression for T c is where Y u and Y v are the parallel admittances in u and v directions. Fig. 4 shows the simulated configurations of the PCM 1 and its mirror unit. In the simulation process, the commercial software CST is used to analyze the properties of these unit cells. Two waveguide ports are set to solve for the reflection and transmission coefficients; Port 1 and Port 2 are positioned a distance of 15 mm away from the PCM 1 . The magnetic and electric wall boundary conditions are enforced along the ±xand ±y-directions, respectively. Fig. 5 presents the simulated and calculated results of the PCM 1 . It can be observed that the simulated and calculated results coincide with each other.    shows the reflection coefficient of the PCM 1 when the incident wave propagates along the +z direction, with the simulated and calculated reflection amplitudes in black solid and blue dotted lines, the simulated and calculated reflection phases in red dash-dotted and magenta dashed lines. It shows that the PCM 1 has a positive reflection phase gradient band of 9.5-12.4 GHz, and the reflection amplitude is better than −3.7 dB in this band. Fig. 5 (b) presents the reflection and transmission amplitudes of the co-polarized and cross-polarized waves when the x or y polarized wave irradiates the PCM 1 along the -z direction. The simulated results of R yx , R xx , T yx , and T xx are in black solid, red dashed, blue dotted and magenta dash-dotted lines, while the calculated results of R yx , R xx , T yx , and T xx are in square orange dash-dotted, circle olive short dashed, up triangle dark gray short dotted, and down triangle purple short dash-dotted lines. It is illustrated that the reflection amplitude of the cross-polarized wave is better than −3.3 dB in the band of 12-26 GHz, thus the PCM 1 can achieve the polarization conversion of the reflected wave in a wide band. Between 8 GHz and 11 GHz, the reflection amplitude of the co-polarized wave is greater than −7.11 dB. Also, it is shown that the transmission amplitude of the co-polarized wave is better than −3.42 dB over 8-11 GHz, and the transmission amplitude of the cross-polarized wave is below −9.27 dB in the entire frequency band. The simulated results of the mirror unit of the PCM 1 are the same as those of the PCM 1 unit. Therefore, when the incident waves irradiate the CPCM, which consists of the PCM 1 unit and its mirror unit, along the -z direction, part of the waves will be reflected directly and the other parts transmitted into the cavity through the CPCM within the X-band. Concurrently, over 12-26 GHz, most of the incident waves will be reflected, and their polarization directions can be converted.

B. ANALYSIS OF LOW-RCS CHARACTERISTICS
Low-RCS antennas play an important role in defense applications. The RCS reduction mechanism of the FP antenna can be analyzed by ray theory, as shown in Fig. 6. In this design, the CPCM is used, and it is necessary to eliminate the co-polarized as well as the cross-polarized reflected waves of the CPCM. The co-polarized reflected wave can be consumed by the retransmitted wave which is formed by the incident wave transmitted into the FP cavity through the CPCM and retransmitted again after multiple reflections. The phase difference between the reflected waves and the retransmitted waves reflecting n times can be expressed as [45] Dif phase = (n + 1)ϕ t + (n − 1)ϕ r + 2nh c k + nϕ GND (5) where ϕ t and ϕ r represent the transmission and reflection phase of the PCM, respectively; k is the wave vector in free space, and ϕ GND = π is the reflection phase of the metal ground of the source antenna. As the phase difference between the reflected wave and the retransmitted wave is close to 180 • ±30 • , destructive interference occurs between the retransmitted wave and the reflected wave. Fig. 7 (a) shows the phase curves of the co-polarized reflected wave (S 11 ) and co-polarized transmitted wave (S 21 ) when the incident wave irradiates the PCM 1 along -z direction. The phase difference between the reflected waves and the retransmitted waves can be calculated with (5) by means of the results obtained in Fig. 7 (b). It shows the phase difference between reflected waves and the first five retransmitted waves. The green region indicates a phase difference of 180 • ±30 • . The calculated results show that, over 8-10.9 GHz, the phase difference between the reflected waves and retransmitted waves is in the range of 180 • ±30 • , thus destructive interference occurs between the retransmitted waves and the reflected waves to reduce the RCS of the FP antenna.
In addition, the PCM 1 and its mirror unit are etched on the dielectric substrate with chessboard arrangement to eliminate the reflected waves after the polarization direction of the reflected wave is converted. The reflection phases of the two units and the phase difference between the reflected waves, as the incident waves irradiate the PCM 1 and its mirror along the -z direction, are presented in Fig. 8. It shows that the PCM 1 and its mirror have perfect phase difference of 180 • in a wide band. Therefore, over 12-26 GHz, the phase difference of reflected waves of two PCM units conforms to the principle of destructive interference so that the wideband RCS reduction of the FP antenna can be obtained due to the destructive interference of reflected waves.
The desired FP antenna operates in X-band. From the aforementioned analysis, the PCM 1 has high out-of-band polarization conversion efficiency. It can be found that the in-band RCS reduction is mainly attributed to the destructive interference between retransmitted and reflected waves, and the out-of-band RCS reduction is mainly determined by the polarization conversion properties of the PCM 1 and its mirror. Consequently, the CPCM, which consists of the PCM 1 unit and its mirror arranged in a chessboard, can reduce the in-band and out-of-band RCS over a wide band.   Table 1. Fig. 10 shows the simulated results of the primary antenna which is seen to have a 10-dB impedance bandwidth of 8.99-11.55 GHz (24.9%), with a maximum realized gain of 9.14 dBi at 11.1 GHz. Its operating band VOLUME 8, 2020  The designed PCM 1 unit and its mirror are etched on the F4BM265 dielectric substrate (ε r2 = 2.65, tan δ = 0.0007) with a thickness of h 1 = 3 mm and a width of W = 80 mm in chessboard arrangement to form the CPCM. The CPCM is placed a distance h c above the primary antenna, as shown in Fig. 11. The CPCM is composed of 3 × 3 tiles, and each tile consists of 4 × 4 PCM 1 units or its mirror units. The electromagnetic waves radiated by the primary antenna produce multiple reflections in the cavity and transmit through the CPCM along the +z direction. When the height of the cavity and the reflection phase of the CPCM satisfy (1), the FP cavity resonance will be excited.
The cavity height has an important influence on the bandwidth and gain of the FP antenna. Fig. 12 and Table 2 show the simulated results of |S 11 | and the realized gain with different cavity heights. Table 2 shows that, as height h c increases from 16.5 mm to 18.5 mm, the 10-dB return-loss bandwidth increases, and the corresponding 3-dB gain bandwidth  expands, while the maximum gain and GBP are increased first and then decreased. After comprehensive consideration, h c = 17.5 mm is selected. In this case, the FP antenna has a 10-dB return-loss bandwidth of 8.65-12.05 GHz (32.9%), a peak directivity of 17.28 dBi at 9.7 GHz, and a GBP of 1422.
From the above analysis, the 10-dB return-loss bandwidth of the antenna expands from 24.9% to 32.9%, and the maximum realized gain increases from 9.14 dBi (11.1 GHz) to 17.09 dBi (9.7 GHz). The bandwidth expansion of the antenna is attributed to the introduction of two additional resonance peaks after the CPCM loading. Meanwhile, the gain enhancement can be explained by (2). For example, at 9.7 GHz, the directivity of the primary antenna is D p = 8.75 dBi, the reflection amplitude of the PCM1 is R = 0.74, so the directivity of the FP antenna can be calculated as D = 17.05 dBi which agrees well with the simulated directivity. In addition, as shown in Fig. 13, the FP antenna has a wideband RCS reduction property over 8-26 GHz, and a peak RCS reduction of 18.06 dB at 11.5 GHz.

C. FABRY-PEROT ANTENNA WITH EMBEDDED CPCM
The X-band was widely used in stealth radars, such as airborne radar [50], [51], shipborne radar [52], [53], and air defense radar [54]. Its relative short wavelength enables higher resolution imaging for target identification and discrimination [55], [56]. For better stealth, improving the in-band RCS reduction of the designed FP antenna is an important issue. From Section II, the in-band RCS reduction is mainly caused by the destructive interference between the retransmitted and reflected waves. As shown in Fig. 5, in the 8-12 GHz range, the co-polarized reflection amplitude of PCM 1 towards lower frequencies is larger than that of the co-polarized transmission, while the co-polarized reflection amplitude towards higher frequencies is smaller than that of the co-polarized transmission. In order to improve the in-band RCS reduction of the FP antenna, it is necessary to improve the in-band energy balance of the co-polarized reflected and co-polarized transmitted waves of the CPCM. For this purpose, the PCM 2 unit and the embedded CPCM are designed, as presented in Fig. 14 and Fig. 15. Due to the consideration of reducing the effect of the small CPCM on the FP antenna performance, a small CPCM replaces the central tile of the original CPCM to form the embedded CPCM. The small CPCM is composed of 2×2 small tiles, and each small tile consists of 2×2 PCM 2 units or its mirror units. The optimized parameters of the PCM 2 unit are as follows: p = 4.8 mm, r 2 = 1.6 mm, r 3 = 2.3 mm, w 2 = 0.6 mm, w 3 = 0.1 mm. Fig. 16 shows the simulated results of the reflection and transmission coefficients of the PCM 2 . Fig. 16 (a) presents the reflection coefficient when incident waves propagate along the +z direction, with the reflection amplitude in black solid line and reflection phase in red dash-dotted line. It shows that the PCM 2 has a positive reflection phase gradient characteristic in the band of 8.4-9.4 GHz, and the reflection amplitude is greater than -8.97 dB. Compared with the PCM 1 , the positive reflection phase gradient band of the   PCM 2 moves towards the low-frequency direction, and the reflection amplitude decreases in the low-frequency band. Obviously, around 10 GHz, the reflection amplitude of the embedded CPCM is larger than that of the CPCM, which contributes to improving the gain enhancement. Fig. 16 (b) shows the reflection and transmission amplitudes of the co-polarized and cross-polarized wave when the linearly polarized wave irradiates the PCM 2 along the -z direction, R yx in black solid, R xx in the red dashed, T yx in blue dotted and T xx in magenta dash-dotted lines. It can be seen that the reflection VOLUME 8, 2020  amplitude of the cross-polarized with reflected wave of the PCM 2 is better than -3.33 dB in the band of 12-22.93 GHz. In the 8-12 GHz range, the co-polarized reflection amplitude of the PCM 2 towards lower frequencies is smaller than that of the co-polarized transmission, while the co-polarized reflection amplitude towards higher frequencies is larger than that of the co-polarized transmission. After embedding the small CPCM into the CPCM, the energy of co-polarized reflection and co-polarized transmission of the embedded CPCM are more balanced, so that the retransmitted waves and reflected waves cancel each other more thoroughly.
The performance of the FP antenna with the embedded CPCM is investigated. Fig. 17 shows the monostatic RCS of the improved FP antenna. It can be seen that the RCS reduction of the improved FP antenna has been significantly improved in the band of 8.0-10.1 GHz; the maximum RCS reduction is 21.06 dB at 9.7 GHz, while the RCS reduction of the antenna decreases in the band of 10.1-11.1 GHz. This is essentially due to the increase of the transmission   amplitude of the co-polarized wave around 9 GHz and the decrease of the transmission amplitude around 10.5 GHz when the incident waves irradiate the embedded CPCM along the -z direction. Also, the simulated RCS result shows that the  impact of the embedded CPCM on the out-of-band RCS can be neglected.
The simulated results of |S 11 | and the realized gain of the improved FP antenna are displayed in Fig. 18 where Fig. 18 (a) and Fig. 18 (b) show that the improved FP antenna has a 10-dB return-loss bandwidth of 8.55-12.11 GHz (34.5%), a 3-dB gain bandwidth of 8.95-11.54 GHz (25.3%), a maximum realized gain of 17.31 dBi at 10.2 GHz and a GBP of 1362. Compared with the primary FP antenna, the improved FP antenna has a wider 10-dB return-loss bandwidth and a larger maximum realized gain, but the 3-dB gain bandwidth decreases, so the GBP decreases. The 10-dB returnloss bandwidth expansion of the improved FP antenna is mainly due to the higher Q value that can be obtained by the centrally embedded small CPCM. The narrower 3-dB gain bandwidth of the improved FP antenna is caused by the fact that the gain becomes lower in the low-frequency band and higher in the high-frequency band. This is principally due to the centrally embedded small CPCM which improves the transmission amplitude of the CPCM in the low-frequency operating band and enhances the reflection amplitude of the CPCM in the high-frequency operating band. Moreover, the reflection amplitude of the embedded CPCM is larger than that of the original CPCM around 10 GHz, which makes the embedded CPCM contribute more to gain improvement.

IV. EXPERIMENTAL RESULTS
The FP antenna with embedded CPCM has been fabricated and measured. The prototype consists of a slot-coupled patch antenna and a substrate etched with the embedded CPCM. The slot-coupled patch antenna is etched on an F4BM220 substrate with thickness of 0.8 mm, and the embedded CPCM is etched on an F4BM265 dielectric substrate with thickness of 3 mm, length and width of W = 80 mm. The overall size of the fabricated antenna is 80 mm × 80 mm × 21.3 mm. All measurements were performed in a microwave anechoic chamber as shown in Fig. 19. Fig. 20 shows the measured results of |S 11 | and realized gain for the improved FP antenna. The improved FP antenna has a measured 10-dB impedance bandwidth of 8.48-12.21 GHz (36.1%), a measured 3-dB gain bandwidth of 8.9-11.5 GHz (25.5%) and a measured maximum realized gain of 17.2 dBi at 10.0 GHz. These values agree very well with simulations. The measured monostatic RCS of the improved FP antenna is shown in Fig. 21. It shows that VOLUME 8, 2020 the FP antenna with embedded CPCM has an RCS reduction band of 8-26 GHz and a peak RCS reduction value of 21.2 dB at 9.4 GHz.
The simulated and measured radiation patterns at 9 GHz, 10 GHz and 11 GHz are depicted in Fig. 22. These patterns show that the improved FP antenna has good radiation directivity. Also, the half-power beam widths (HPBWs) of the FP antenna in the E-plane and H-plane at 10.0 GHz are 21.5 • and 21.7 • , respectively. The measurements confirm the correctness of this design. Table 3 exhibits the performances comparison between the proposed antenna and related reports. It illustrates that the FP antenna with embedded CPCM has excellent performance, including a wider 10-dB return-loss bandwidth, a larger gain enhancement, a wider RCS reduction band and a higher GBP.

V. CONCLUSION
A high gain-bandwidth product Fabry-Perot antenna with wideband RCS reduction property is proposed using the embedded CPCM. An FP antenna with CPCM is designed and investigated. The mechanisms of the in-band and outof-band RCS reductions are analyzed. They reveal that the in-band RCS reduction of the FP antenna is mainly attributed to the destructive interference between retransmitted waves and reflected waves, and the out-of-band RCS reduction is mainly determined by the polarization conversion characteristics of the CPCM. The high GBP depends on the FP resonant cavity. Moreover, an additional small CPCM replaces the central tile of the original CPCM to form the embedded CPCM. The embedded CPCM improves the in-band RCS reduction for the FP antenna. Also, it has a positive effect on gain enhancement and 10-dB impedance bandwidth expansion. Experiments show that the FP antenna with the embedded CPCM has a relative 10-dB return-loss bandwidth of 36.1%, a GBP of 1338, a 3-dB gain bandwidth of 25.5%, and a maximum gain of 17.2 dBi at 10 GHz. Meanwhile, the measured RCS reduction band of the proposed FP antenna covers 8-26 GHz (105.9%), with a peak RCS reduction of 21.2 dB at 9.4 GHz. XIANGKUN KONG (Member, IEEE) received the Ph.D. degree in communication and information systems from the Nanjing University of Aeronautics and Astronautics (NUAA), in 2015. He has been an Associate Professor with NUAA, since his promotion in July 2015. He used to work at the University of St Andrews, U.K., as an Academic Visitor supported by the China Scholarship Council. His main research interests include the electromagnetic properties of metamaterials, metamaterials and metasurface application, plasma photonic crystal, and computational electromagnetics. He has published more than 100 articles in different academic journals, including Applied Physics Letters, Optics Express, and the IEEE JOURNAL OF SELECTED TOPICS IN QUANTUM ELECTRONICS. He has been cited 1600 times.
ZHENGYU HUANG was born in Hunan, China, in 1986. He received the B.S. degree in electronic and information engineering from Nanchang University, Nanchang, China, in 2008, and the Ph.D. degree in electromagnetic environmental effects and electro-optical engineering from the PLA University of Science and Technology, Nanjing, China, in 2016.
He is currently an Associated Professor with the Key Laboratory of Radar Imaging and Microwave Photonics, Nanjing University of Aeronautics and Astronautics, Nanjing. He has coauthored one monograph on the Associated Hermite finite-difference time-domain method and 30 peer-reviewed journal articles included in the Web of Science Core Collection. His research interests include computational electromagnetics, electromagnetic compatibility, and multiphysics modeling and analysis. He was a recipient of the Outstanding Doctoral Dissertation Award of the PLA University of Science and Technology, in 2017, and the Excellent Doctoral Dissertation Award by Lu Zhitao Academician, in 2018.